A filtering technique with different configurations is employed to get better isolation. C-shaped filtering stubs as shown in Figure 10a are introduced in [15] for achieving portto-port isolation >25. The purpose of the filtering stub is to control the current flow across the feeding line of the relevant port to act as a band stop for the specific band to achieve high isolation. Filtering response through parasitic elements as shown in Figure 10b is realized in [20], resulting in improved isolation of greater than 32 dB. The basic filtering structures include metasurface structure [46], slot [47], shorting vias [48], and parasitic elements [49], and defected ground structure [50] integrated with the radiator to achieve the filtering response for base station application. Although an extra filtering structure increases insertion loss, the filtering antenna as array elements without extra decoupling structure has been proposed in [24]. The radiating element realized the filtering response by adding the shorting pins and E-slot to achieve inter-band isolation of 35 dB. Balun design is modified to provide required filtering removing the need to have additional filters in [51]. Intra-band isolation >31 dB is achieved in [52] by carefully designing the dipoleto mutual coupling to complement the cross-polar isolation.

A configuration named lower-band–ground–upper-band (L–G–U), where the highband antennas are located above the lower-band antenna separated by a low pass surface, is presented in [53]. It demonstrates inter-band isolation better than 30 dB in both working bands. In [54], a frequency selective surface is introduced between the high band and low band elements in the L–G–U configuration to achieve inter-band isolation >25 dB. The frequency-selective surface is optimized to serve as top capacitive loading for low-band 0.69–0.96 GHz and act as a reflector for high-band 3.5–4.9 GHz. In [55], beam-to-beam isolation > 32 dB was achieved through Luneburg lens. This Luneburg lens antenna operates from 1710–2690 MHz, made up of a special periodic structure to become suitable for base station application. In the literature, a Luneburg lens with different materials was

designed such as in [56], which is configured with a metamaterial layer to make it compact at lower frequencies (0.8–6 GHz).

**Figure 10.** (**a**) Proposed antenna configuration [15]. (**b**) Parasitic element configuration in proposed antenna [20].

#### *3.3. Radiation Patterns*

The far-field radiation patterns are a very important, if not the most important, factor for an operator. Some critical parameters in patterns include 3 dB beamwidth, 10 dB beamwidth, beam squint, front-to-back (F/B) ratio, sidelobe levels, and cross-polarization discrimination (XPD). Some of these parameters are marked in Figure 11. The specifications provided by the operators to antenna designers can slightly change but mostly follow the industry-standard requirements listed in [34].

**Figure 11.** Illustration of some critical parameters for antenna radiation pattern.

The beamwidths 3 dB and 10 dB are important in network planning to mark sector footprints. Usually, single beam/sector antennas must have an HPBW requirement of 65◦ for three sector cell sites and 45◦ or 33◦ for six sector sites. The 10 dB beamwidth refers to the angular beamwidth at 10 dB below the peak values. It is desirable to have a 10 dB beamwidth of 120◦ in a three-sector site which implies that signal strength at the sector edge is 10 dB below the peak in boresight.

The radiation element most often needs modifications in its design to achieve the desired azimuth beamwidth. The important factors that affect the azimuth beamwidth are the length of the radiator and the distance between the radiator and reflector. These parameters are optimized to attain 3 dB beamwidth within the range of 65.7◦ ± 3.2◦ [19]. The use of dipole-type radiators is common in radiating element design. The umbrellashaped dipoles are configured in [57] to achieve an HPBW (half-power beamwidth) of 63◦ ± 5◦ in the H-plane. Another technique to modify the azimuth beamwidth is the use

of cavity-shaped reflector over a planer reflector [58]. Further, rectangular cavity-shaped reflectors in [16] and convex-shaped reflectors in [19] are used to achieve 3 dB beamwidth around 65◦ ± 5◦. The length and height of the reflector primarily affect the 3 dB beamwidth in the H-plane. This effect is realized in [10] to achieve a 3 dB beamwidth 68◦ ± 2◦ at Hplane and V-plane. It is found that increasing the length of the box-shaped reflector narrows the beamwidth in lower frequencies and widens the beamwidth in high frequencies in the band 1.7 to 2.7 GHz.

Another critical parameter in radiation patterns is the front-to-back (F/B) ratio. A higher F/B ratio is desirable to minimize backward radiation that can contribute to cochannel interference. Typical values need to be 25 dB or more in most of the base station antenna designs. Having a large reflector helps improve the F/B ratio. However, most of the BSAs have size constraints in terms of antenna width. Different topologies have been adopted in the literature to achieve the desired F/B ratio. Radiating elements proposed in [25] have downward-sloping dipoles, which improves the F/B ratio. An F/B ratio better than 30 dB and sidelobe levels better than 25 dB were achieved. Placing a large reflector behind BSA can improve F/B; however, it can introduce other intricacies such as increased wind resistance, large antenna size, and antenna loading. An electromagnetic scattering structure applied on the Radome is used in [59] to reduce back lobe radiation.

Beam Squint is another critical performance parameter for antenna radiation patterns. The beam squint refers to the deviation of the main beam direction from its boresight in the azimuth plane. The beam squint is measured in ±degrees from the boresight direction. The squint can be measured as a 3 dB beam squint or a 10 dB beam squint, and the aim is to keep the squint as low as possible throughout the entire operating band for all elevation tilts. Generally, the squint gets worse with higher elevation tilts. In [26], beam squint up to 12◦ is reported at 10◦ elevation tilt. Different techniques are followed to minimize the beam squint. In [27], a combination of microstrip and a stripline PCB's are introduced in the radiation element to minimize the beam squint below 5◦ for downtilt measured at 7◦ and 0◦. Another technique is to enforce symmetric current distribution on radiating elements to minimize squint. This is achieved in [38] by using the Pawsey stub balun to feed radiating elements and to reduce leaky current distortions to have HPBW within 54–76◦. In [28], the beam squint < 4◦ with a maximum 10◦ down tilt is achieved through octagon-shape-folded dipoles as shown in Figure 12a.

**Figure 12.** (**a**) Fabricated prototype of folded dipole [28]. (**b**) Fabricated Prototype of magnetoelectric loop dipole [29].

Cross-polar discrimination (XPD) is another important parameter when looking at radiation pattern performance. In general, the XPD above 10 dB in the sector is recommended for base station application. The magnetoelectric layered loop dipole configuration shown in Figure 12b, is used in [29] to achieve XPD over 20 dB in the boresight direction. Antenna gain is also a vital characteristic in BSA design. Operators prefer a positive gain slope across the band as the higher gain at higher frequencies compensates for the additional

free space losses. In the azimuth, the gain in sector edge directions is lower compared to boresight due to gain roll-off. A lower gain roll-off in azimuth patterns is achieved in [30] by introducing a notch metal wall to radiating element, which enhances the gain by 2 dB at sector edge ± 60◦ angle.

The aforementioned radiation pattern-related parameters are applicable for singleband and multiband antennas. However, in multiband antennas, radiation patterns can be severely impacted due to cross-band scattering when interleaved or embedded arrays. In a dual-band interspaced array, the high-frequency element patterns are impacted due to scattering from the currents induced in low-frequency elements and vice versa. The impact can be on multiple parameters such as beamwidth, squint, and XPD. Overcoming such impacts can be very difficult. It is often attempted to ensure that each radiating element is transparent to the other in their operating frequency bands, which is a challenging task. In [60], the high-frequency band (HB) pattern distortions caused by the lower frequency band (LB) radiating elements are minimized by introducing chokes into the LB element. These chokes are quarter-wavelength open circuit segments at high band frequencies, and this minimizes scattering. In [61], the printed dipole is segmented into smaller segments that are not resonant in the higher frequency band region, and each segment is connected to the other by inductive thin lines. This makes the lower band element transparent to the higher band radiating element. In [62], a cloaked antenna system is realized to minimize the scattering of closely located antennas. A dual-polarized mantle cover to cloak the dipole antenna is used in this work, and radiation performance is almost unaffected.

The aforementioned challenges for patterns are fundamental and equally applicable for single beam, multibeam, or steerable beams in BSAs. When it comes to 5G, there are some additional challenges and complexities associated with the mMIMO and beamforming. As mentioned earlier in Section 2, there are two main beamforming technologies, namely analog and digital, used in BSAs for sub-6 GHz. Although the analog beamforming is not true mMIMO, it is still used in some of the 5G base station antennas to form multiple beams. The true mMIMO in sub-6 GHz is achieved through digital beamforming in 5G BSAs.

The most popular techniques used for analog beamforming are based on either Butlermatrix circuits [63] or Luneburg lenses [64]. Compared to lenses, the Butler-matrix circuit implementations are compact, low-cost, and planar. The designers can incorporate the Butler matrix implementations with the feed distribution network. Therefore, it does not necessarily increase the antenna height. However, there are several challenges associated with the Butler matrix implementation such as dual-band operation, isolation between beams, side-lobe suppression, and wide operating bandwidth [65]. The branch line couplers used in the Butler matrix have inherent bandwidth limitations, and as a result, they cannot be designed to have multi-band operation. The approach is to have distinct Butler matrices for each band [66] to overcome this limitation. However, this comes with inherent crossband coupling challenges, which were discussed in Section 3.2. The narrowband challenges were addressed with wideband quadrature couplers and fixed-phase shifters in a Butler matrix [67]. Although there has been some reported literature on high-beam-to-beam isolation [68], it is still an ongoing challenge. One solution to minimize the sidelobes as well as grating lobes is achieved by changing the antenna element arrangement in [63].

In contrast to butler matrices, a careful design of a lens-based beamformer can provide stable radiation patterns, with low sidelobe levels in a wide band [8]. However, the size of the lens in front of the antenna increases the antenna height as well as weight. Therefore, low profile, low cost, and lightweight lenses remain a potential research topic for analog beamforming in BSAs. The other analog beamforming techniques reported in the literature include the use of metasurfaces [69], parabolic cylindrical reflectors [70], and reconfigurable parasitic radiators [71].

The digital beamforming architecture, which provides true mMIMO capability, has certain challenges as well. One of the key challenges is the design complexity. An example 4 × 4 Tx/Rx dual polar mMIMO configuration for a BSA is shown in Figure 13. Each

radiating element requires an RFIC to provide amplification and filtering, which are then connected to the digital beamformer. This dual polar Tx/Rx array requires in total 32 RFICs and 8 baseband beamformers. All these components need to be placed closer to radiating element, making the integration of the antenna element and RF circuitry very complex. Unlike mm-wave designs, the sub-6 GHz designs cannot be realized as an integrated AiP in MMIC technology due to the large size of the antenna element. In addition, the large number of RFICs and digital processing results in high power consumption. Even with the state-of-the-art efficient power amplifiers available in the sub-6 GHz bands, the thermal dissipation from processors and RFICs remains a significant challenge that needs to be addressed in the design. Thermal vias and thermal pads are necessary at the back of the RF and digital electronics to dissipate the heat in these antennas [31]. The increased power consumption results in less value for money for the operators despite the capacity improvement. Some antenna designs have addressed this by limiting the real-time beamforming on the horizontal axis only [64]. The elevation beam tilting in this case is not done electronically and initially set by the remote electrical tilting mechanism using the legacy phase shifters. Then, the Azimuth beamforming is done using the phase and amplitude control among array columns.

**Figure 13.** Components of a 4 × 4 Tx/Rx dual polar mMIMO configuration.

Another challenge with the mMIMO antenna design is the calibration. The phase needs to be calibrated with high accuracy along the entire RF transceiver chain in order to ensure the expected beamforming gains. The amplifiers can have varying phases based on their operating conditions such as bias points, temperature, and frequency of operation, and these variations can lead to deviations from the expected phase distribution at the elements. As a result, the patterns may not form nulls, where it is expected that leads to increased interference with adjacent users. The antenna testing is another challenge for mMIMO as the access to RF inputs of the antenna is difficult with the other components in the RF transceiver chain. Therefore, instead of doing antenna pattern testing with an RF input, Over-the-Air (OTA) testing is required with the baseband IQ data as the input/output.

#### *3.4. Size and Cost*

Operators are keen to maximize the performance per unit area in the tower space. Therefore, they require compact antennas with better overall performance at a low cost. This requires the designs to be compact and low in cost. The miniaturizations are often achieved by having interspersed multiband arrays [4]. A comparison of different arrangement schemes for high-frequency elements and low-frequency elements is performed [2]. As shown in Figure 14, side-by-side schemes, up-and-down coaxial schemes, and embedded schemes were considered. Although the up-and-down scheme is simpler, cable losses increase due to the increased length of the main feed line. The embedded scheme has the advantage of compactness as two antennas are located within a single band antenna volume. However, embedded schemes exhibit challenges of isolation and pattern distortions. A single radiating element for both the low and high bands is used in[72,73] to reduce the number of required radiating elements and footprint. However, the required vertical element spacing for optimum elevation patterns is hard to achieve in this approach. Hence, the grating lobe levels are high. The dual broadband planer BSA configuration is followed in [32], where high band elements are nested inside lower band elements to achieve compactness.

**Figure 14.** (**a**) Multiband embedded scheme; (**b**) multi band side-by-side scheme; (**c**) multiband -up-and-down coaxial scheme [2].

The lower cost in the antenna production is achieved by using low-cost, simple, printed antennas. The printed circuit board technologies reduces antenna assembly times. Another low-cost choice for enabling 5G BSA is 3D printing, which facilitates complex designs. Three-dimensional printing is an effective manufacturing method for designing MIMO antenna prototypes to reduce the cost. In [33], ±45 dual-polarized antenna is fabricated using 3D printing technology while achieving a wide bandwidth. Although printed antennas have a low-cost advantage, they can exhibit higher dielectric losses compared to die-cast radiating elements. Die-casting can be cost-effective if the entire element is made out from a cast reducing the assembly time. In addition to the cost incurred on radiating elements, the other RF components, such as phase shifters, may incur higher costs. A wiper phase shifter [74] can be a cost-effective solution as it is very compact but may have other disadvantages, such as limited control for null-filling in down tilts and the use of a large number of cables. Other types of phase shifters such as [75,76] can overcome some of these limitations while reducing fabrication costs.

#### **4. Future Trends in 5G Base Station Antennas**

The foundation of any BSA antenna is the radiating elements in the arrays. Therefore, it is important to select wideband radiating elements with stable radiation patterns. Table 4 contains some of the recent state-of-the-art radiating elements that show wideband performance. Some of the design concepts used in the design of these elements can be used as the basis to develop more improved radiating elements for current and future 5G antennas.

*Electronics* **2021**, *10*, 2000

size in terms of wavelength(λ) at midband frequency, # Not Given.

\*

The massive MIMO is one of the most popular topics among the 5G BSA community. Currently, the massive MIMO used in the 5G network is typically standalone, i.e., contains only one frequency band commonly referred to as mid-band (2.6 GHz–4.2 GHz) with 32 or 64 ports (eight dual-polarized columns with two or four rows). No other bands are integrated into those active antennas. The massive MIMO antennas require the radios to be used for each of the ports. Despite the high throughput advantage of these massive MIMO antennas, the extreme power consumption puts a significant loading on the electrical network. The operators cannot remove the existing 4G BSAs to reduce the power consumption since the current 5G massive MIMO antennas cannot still serve all the frequency bands. Therefore, the industry is leaning towards integrating legacy multiband 4G antennas with 5G massive MIMO antennas. One potential approach is to reduce the number of ports in massive MIMO BSAs from 32 to 16 and provide some space to integrate a legacy antenna. The aim is to strike a balance with operational cost and performance. The combination of legacy 4G and 5G massive MIMO still poses the challenges highlighted in the paper. For example, the low band (under 1 GHz) needs to be decoupled in an even wider band from 1.4 GHzto 4.2 GHz, which is very difficult. Moreover, the antenna profile is required to be slimmer for less wind loading, making the technical design even harder. Possibilities remain open to design novel wideband frequency selective surfaces to overcome some of these challenges to reduce coupling and improve patterns since more bands are now required to be integrated under one radome.

#### **5. Conclusions**

Base Station Antennas have evolved from simpler Omni antennas to multiband multibeam sectorized antennas over the last three decades. The antenna complexity has increased with each wireless generation, leading to more design challenges to antenna engineers. From LTE-A leading to 5G, sub-6 GHz BSA designs have multiple antenna arrays. This is due to several reasons, including the need to support MIMO capability and the need to maximize value for money by covering multiple bands. To achieve size savings, the multiple bands are interspersed in the design. The presence of multiple radiators introduces scattering, degrading the pattern performance of each band. Coupling between the close proximity elements leads to challenges in meeting impedance and isolation specifications. As a result, multiple techniques are applied by antenna designers such as cloaked dipoles, parasitic elements, FSSs, and filters to mitigate these challenges.

**Author Contributions:** All authors conceived and proposed the idea; all authors contributed to the structure of the paper. Wrote the paper under the supervision of D.N.T. All authors have read and agreed to the published version of the manuscript.

**Funding:** This research received no external funding.

**Data Availability Statement:** The data presented in this article can be obtained from the relevant references provided in this article.

**Conflicts of Interest:** The authors declare no conflict of interest.

#### **References**

## *Article* **A Sub-6 GHz MIMO Antenna Array for 5G Wireless Terminals**

**Jalal Khan 1, Sadiq Ullah 1,\*, Farooq A. Tahir 2, Faisel Tubbal 3,4 and Raad Raad <sup>3</sup>**

**Abstract:** This paper presents a novel antenna with its array and MIMO configuration for the 5G sub-6 GHz applications. The proposed antenna element operates at the central frequency of 5.57 GHz dedicated for Sub-6 GHz 5G communication applications. The antenna element holds a circularshaped radiating portion with an inner-circular slot, plus a rectangular slot at its right edge to make the proposed design resonate at the desired frequency band. The RT5880 substrate is used with a thickness of 0.787 mm, and the low-loss tangent of 0.0009. To achieve a desired gain of 12 dB, a four-element array configuration is adopted, which improved a bore side gain to 12.4 dB from 6.66 dB. Then, the two-port configuration is adopted such that the isolation achieved between them is more than −30 dB. The total efficiency of the proposed antenna array is observed to be more than 80% within the operating bandwidth. Moreover, the Specific Absorption Rate (SAR) analysis is also presented for the proposed MIMO configuration, obeying the standard value (i.e., <2 W/kg for any 10 g of tissue). The measured results are in good agreement with the simulated results. All the simulations of the proposed design are performed in the CST MWS software.

**Keywords:** 5G; antenna array; CST; sub-6 GHz; MIMO; SAR

#### **1. Introduction**

The day-by-day increase in the demand for higher data rates and bandwidth has created several issues for the current fourth-generation wireless communication infrastructure [1], as the current wireless communication infrastructure lacks the potential to deliver a higher data rate with a low latency [2]. Thus, the fifth generation has been introduced, which is promising to fulfil the requirement for higher data rates with the desired low latency level [3].

At the moment, most of the designs reported are single element, which possess quite low gain as compared to the desired level (>12 dB) required for 5G communication [4–8]. Likewise, several multiple-port antennas have been presented to improve data rates, but they could not achieve satisfactory gain [9–13]. In the future, these Sub-6 GHz 5G antennas will be integrated with their predecessors and some other higher-band application antennas on the same printed circuit board (PCB). Thus, the high-level of gain can be quite helpful to efficiently deliver a stronger signal at the user side, instead of congesting the lower frequencies of the spectrum [14]. Along with this, the multiple input multiple output (MIMO) configuration is quite helpful to provide a good channel capacity and data rate as compared to the single port antennas, which is one of the basic challenges in 5G transmission.

Recently, some MIMO antennas have been reported for operation in the Sub-6 GHz band [15–20]. Different configurations have been adopted—such as monopole, open-slot,

**Citation:** Khan, J.; Ullah, S.; Tahir, F.A.; Tubbal, F.; Raad, R. A Sub-6 GHz MIMO Antenna Array for 5G Wireless Terminals. *Electronics* **2021**, *10*, 3062. https://doi.org/10.3390/ electronics10243062

Academic Editor: Christos J. Bouras

Received: 5 November 2021 Accepted: 5 December 2021 Published: 9 December 2021

**Publisher's Note:** MDPI stays neutral with regard to jurisdictional claims in published maps and institutional affiliations.

**Copyright:** © 2021 by the authors. Licensee MDPI, Basel, Switzerland. This article is an open access article distributed under the terms and conditions of the Creative Commons Attribution (CC BY) license (https:// creativecommons.org/licenses/by/ 4.0/).

or loop antenna design method to achieve operation in the desired frequency band. As in [15], the size of the proposed design noted is 130 mm × 100 mm with an operating band of 5.15–5.925 GHz. The gain obtained ranges from 2.5–4.2 dB within the operating band. Similarly, another antenna with an overall size of 150 mm × 75 mm and operational bandwidth of 5.15–5.85 GHz is presented in [16]. The peak gain noted is 4.62 dB. Moreover, an antenna with a size of 136 mm × 68 mm is presented in [17]. The proposed design adopted a monopole configuration in order to operate in the band of 5.15–5.925 GHz, while the gain achieved is not discussed. Apart from this, the main drawback observed is a low efficiency in [18–20], while in [21–23], the efficiency attained is even below 50%. In [24], the efficiency obtained ranges between 50–82%, while the antenna element is extended to the multi-port configuration over the same PCB with an overall size of 150 mm × 75 mm. However, the minimum isolation level observed is −12 dB, which can be a great cause of signal deterioration on the user side.

In this paper, we have presented a microstrip circular patch antenna array designed at the 5.5 GHz band with four elements in the array structure using microstrip methods, and then its MIMO configuration for 5G communication applications. The transmission line and quarter wavelength transformation techniques are used for the impedance and phase matching. Moreover, the SAR analysis is also presented for the proposed MIMO configuration. The single element, four-element array, and MIMO antennas are analyzed using CST Microwave Studio. The remainder of the paper is organized as follows: Section 2 presents the design methodology of the proposed antenna. The results and discussion have been presented in Section 3, and Section 4 discusses the SAR analysis. Section 5 concludes the paper and gives recommendations for the future.

#### **2. Antenna Geometry**

#### *2.1. Single Element*

The front view of the antenna element is shown in Figure 1. RT5880 with a thickness of 0.787 mm is used as a substrate. The length and width of the substrate is 50 × 40 mm2. The radiating portion of the antenna exhibits a circular, patch-like geometry, with a circularshaped slot in the center with an inner radius of 6.75 mm and an outer radius of 11.54 mm. A slot-termed gap is created at the edge of the patch to tune the resonant frequency to the desired frequency band. The feed contains an impedance transformer for the purpose of matching the transmission line with the radiating portion of the antenna. The back side of the proposed antenna is fully copper-coated. The parameters in Table 1 have been calculated via the known equations of the transmission line theory for the circular patch [25] and have then been optimized to achieve resonance at the desired frequency band.

$$\text{rad}\_1 = \frac{\text{F}}{\left\{1 + \frac{2l}{\pi \varepsilon\_r \text{F}} \left[\ln\left(\frac{\pi \text{F}}{2l}\right) + 1.7726\right]\right\}^{\frac{1}{2}}} \tag{1}$$

$$\text{F} = \frac{8.791 \times 10^9}{f\_r \sqrt{\varepsilon\_r}} \tag{2}$$

*fr*

**Figure 1.** Front view of the antenna element.

The voltage standing wave ratio (VSWR) and reflection coefficient graphs are shown in Figure 2. It is seen that the magnitude of the reflection coefficient is below 15 dB, which is acceptable for optimum power transmission. The VSWR is also observed below 2 dB within the operating bandwidth, which depicts low loss and a good matching between the feedline and the radiating portion of the proposed antenna. The −10 dB bandwidth of almost 50 MHz is obtained with a range from 5.55 to 5.6 GHz, with the central frequency noted to be 5.57 GHz, which is dedicated for 5G Sub-6 GHz applications [18].

**Figure 2.** Antenna element: (**a**) reflection coefficient; (**b**) VSWR.

The parametric study for the proposed antenna element by varying the values of different key parameters is shown in Figure 3. It is observed that the value of the parameters' variations has a significant effect on the reflection coefficient of the proposed design. All three parameters, such as R1, R2 and gap are helpful to bring the antenna element to resonate at 5.57 GHz. In the case of R1, the resonance for the selected parameter is at 5.6 GHz, while simultaneously varying R2 and gap parameter with the R1 moves the resonance to 5.57 GHz. Alongside, the matching circuit helps to improve the magnitude of the reflection coefficient. Thus, after careful consideration and analysis, final values are taken, and the antenna element can proceed towards the array concept.

**Figure 3.** Antenna element reflection coefficient analysis by (**a**–**c**) varying different parameters.

The peak gain patterns for the proposed antenna element are presented in Figure 4. The polar and 3D gain patterns are observed in Figure 4a and Figure 4b, respectively. The gain pattern in the 90-degree plane is analyzed, and the side lobe level is observed to

be −13.8 dB, while the magnitude of the main lobe is 6.65 dB. Further, the angular 3 dB beamwidth is seen to be 110.4 degrees. The 0-degree plane gain pattern is with the side lobe level of −24.1 dB, whereas a main lobe magnitude of 5.93 dB with a 3 dB beamwidth is 69.9 degrees. The gain achieved is 6.66 dB as shown in the 3D radiation pattern (Figure 4b) of the proposed antenna. The greater level of side lobe reduction makes the proposed design less lossy, whereas the back lobes level is also very low as shown in the E plane of Figure 4a.

**Figure 4.** Antenna element: (**a**) gain pattern (polar); (**b**) gain pattern (3D).

#### *2.2. Multiple-Element Antenna Array*

The design of single element is extended to another type of the array technique, which is a most famous gain enhancement technique. In Figure 5, the corporate feed array is presented, where each element holding a symmetry is placed at a distance of 0.477λ0 to achieve a particular gain enhancement. The gap between the neighbor patches is optimized step by step to have minimal performance degradation effect on each other, and finally, 0.477λ0 is chosen as an optimized gap where reflection coefficient, gain, and efficiency could be quite satisfactory. The back of the substrate is fully coated with copper material. Some of the design parameters of the corporate feed array have been shown in Table 2.

**Figure 5.** Corporate feed sub-6 GHz array.

The width of the transmission line is calculated by using the following relationship [26].

$$\mathbf{w}\_{\mathbf{z}\_{\mathbb{G}}} = \left(\frac{377}{\mathcal{Z}\_{\mathbb{G}}\sqrt{\varepsilon\_{\mathbb{F}}}} - 2\right) \times \mathbf{h} \tag{3}$$

where *ε*<sup>r</sup> is the relative permittivity and Zc is the characteristic impedance of the transmission line. In the proposed antenna design a 50 Ω transmission line is used.

#### *2.3. MIMO Configuration*

The array design achieved utilizing the corporate feed technique is extended to the MIMO configuration of two ports as shown in Figure 6. The arrays are placed with a 90-degree shift with respect to each other in order to achieve pattern diversity and good isolation among antenna arrays, which can cause issues when extending arrays to MIMO. The dimensions of the MIMO antenna are outlined in Table 3.

**Figure 6.** Corporate feed array MIMO configuration.

**Table 3.** Design parameters of MIMO array antenna.

#### **3. Results and Discussion**

The results obtained through simulation and measurements for the proposed corporate array depicted in Figure 5 are analyzed in this section. In Figure 7, the measurement setup is shown for the proposed corporate array results analysis. As shown in Figure 8, the reflection coefficient obtained for the corporate feed array is showing resonance at the desired band of 5.57 GHz with a magnitude of 25 dB. In case of corporate feed structure, the arrangement of elements is done such that—in terms of spacing between the elementsthe coupling among them should be negligible. The bandwidth covered by the proposed antenna array ranges from 5.54 to 5.59 GHz.

(**a**) (**b**)

**Figure 7.** Proposed antenna array under testing: (**a**) measuring reflection coefficient and (**b**) radiation measurements inside the anechoic chamber.

**Figure 8.** Corporate feed array reflection coefficient.

The measured reflection coefficient follows the simulated one quite efficiently, with an improvement in the magnitude level. Furthermore, a 3 dB offset is observed between the simulated and measured reflection coefficient due to the losses by connecters. The minor shift in the frequency observed is due to fabrication tolerances. The VSWR is also below 2 for the operating bandwidth while at the central frequency, the value of it is observed to be 1.12, which shows that there is minimum reflection of the transmitted power from the radiating structure to the feed, as shown in Figure 9.

**Figure 9.** Corporate feed array VSWR.

The radiation pattern is highly directive, with a gain of 12.4 dB as shown in Figure 10. It can also be seen that the side lobe level is highly reduced in this type of array structure. According to the 0-degree plane of radiation, the side lobe level is −17.1 dB, which is quite satisfactory for 5G communication, and the main lobe direction is towards 0 degree, while the magnitude of the main lobe is found to be 12.6 dB. The angular width (3-dB beamwidth) is achieved at 22 degrees.

**Figure 10.** Corporate feed array gain pattern: (**a**) *E*-plane; (**b**) *H*-plane.

However, in the 90-degree plane of radiation, the main lobe direction is along a 2-degree angle with a side lobe level of −17.1 dB. The angular width (3-dB beamwidth) in this plane is 82.1 degrees, and the main lobe magnitude is 12.6 dB. Furthermore, the back lobe level in this plane is highly reduced, making the antenna perfect for the 5G communication environment. The measured results follow the simulated with a quite good coherence due to the better fabrication and measurement setup calibration.

The 3D gain pattern further helps to understand the above 2D gain pattern, as shown in Figure 11a using the Computer Simulation Technology (CST) software. The radiation and total efficiency within the operating bandwidth are 98.2% and 85.1%, respectively. The current distribution for the corporate feed structure is observed in Figure 11b, which uses the Computer Simulation Technology (CST) software to show that the current is mostly concentrated along the edges of the circular-shaped patch, and feedline while the current is negligible between the patches; this proves that the coupling among the elements is minimized by employing the corporate feeding technique. Thus, unwanted resonances are avoided in the return loss of the proposed antenna array, and a high gain is achieved with the high efficiency. Moreover, the current distribution in Figure 11b of the whole array ensures that the power distribution is equal to each antenna element. The distribution of the electric (E) and magnetic field (H) are shown in Figure 11c and Figure 11d, respectively.

**Figure 11.** (**a**) The 3D gain pattern of the proposed antenna array, (**b**) current distribution, (**c**) E field and (**d**) H field of the proposed array structure.

The reflection coefficient for the proposed MIMO structure is presented in Figure 12a. For port-1, the antenna resonates with the central frequency of 5.57 GHz, gives a bandwidth from 5.59 to 5.54 GHz, and a magnitude of −25 dB. Similarly, for port-2, the antenna resonates with the central frequency of 5.56 GHz, with a reflection coefficient magnitude of −26 dB. The magnitude of the return loss remains below −10 dB, within the resonant bandwidth (5.58 to 5.53 GHz). The isolation for the proposed MIMO antenna is presented and analyzed in Figure 12b for both the ports with respect to each other. The isolation is observed to be greater than 40 dB within the operating bandwidth, which clearly indicates that the isolation among the arrays employed is quite minimum.

**Figure 12.** MIMO: (**a**) reflection coefficient; (**b**) isolation.

The gain patterns are analyzed and presented in Figure 13a in the 0- and 90-degree planes for the port-1. Initially, in the 90-degree plane, the antenna array main lobe direction is located at 2.0 degrees, while a very good side lobe level of −17.1 dB is achieved with a 3-dB beamwidth of 82.1 degrees. The radiation pattern is quite directive, with minimum back lobes and side lobes. The 0-degree plane analysis shows that the main lobe is directed towards the 0-degree angle and the side lobe level is −17.1 dB. Furthermore, a 3-dB beamwidth is 22 degrees. In this particular plane, the radiated beam is also seen to be highly directive with a low level of back lobes.

Similarly, the radiation pattern analysis in the case of port-2 is presented in Figure 13b for the two main planes: E and H. The main lobe direction for the 0-degree plane is along the 355-degree angle, and the side lobe level is −3.1 dB, while the angular width is 83 degrees in this particular plane. Likewise, the direction of the main lobe is along 1.0-degree angle in the 90-degree plane, with an angular width of 22.4 degree. The side lobe level is −15.1 degrees in this plane. Further in-depth analysis of the radiation feature of the proposed MIMO configuration (antenna array employing a corporate feeding technique) is presented in Figure 14. The port-1 or antenna-1 has a narrow and directive beam with a peak gain of 12.6 dB.

**Figure 13.** MIMO polar radiation pattern: (**a**) port-1; (**b**) port-2.

**Figure 14.** MIMO 3D radiation pattern: (**a**) port-1; (**b**) port-2.

The radiation efficiency and the total efficiency is 98.2% and 85.1%, respectively. Furthermore, for port-2 the radiation beam is shifted by a 90-degree angle, which shows

that pattern diversity) is achieved with the proposed MIMO structure. The peak gain of 12.6 dB is achieved with radiation and total efficiency of 97.8 and 79.6%, respectively.

The Envelope Correlation Coefficient (ECC) is another parameter that describes the level of correlation among multiple antennas which come into close proximity of each other. For the proposed MIMO configuration of two port antenna array, the ECC value is below 0.005 for the entire operating bandwidth, which satisfies the standard criteria of a value of <0.5, and depicts that isolation among the antenna arrays is minimum as shown in Figure 15a. Another performance metric of MIMO is Diversity Gain (DG)—meeting or coming close to the standard value of 10 dB shows that reduction in the transmitted power will have no major effect on the quality of transmission or MIMO performance. As seen in Figure 15b, the DG value is close to 10 for the entire operating bandwidth, which satisfies the standard criteria. All the results discussed above show that MIMO-based corporate feed arrays are very suitable for future 5G Sub-6 GHz communication.

**Figure 15.** MIMO: (**a**) ECC; (**b**) DG.

#### **4. SAR Analysis for the Proposed MIMO Antenna**

The SAR analysis for the proposed MIMO array antenna is discussed in this section. The proposed model contains skin, fat and muscle with a thickness of 1 mm, 3 mm and 30 mm, respectively as shown in Figure 16. Skin is used as the topmost layer, while the fat and muscle are placed behind the skin in that order. The proposed MIMO configuration (i.e., corporate feed array) is placed at the center of the body model and the performance is analyzed. A conductivity of 3.5 S/m and relative permittivity of 35.3 have been used for the skin, while for fat and muscle, a conductivity and relative permittivity of 0.27 S/m, 4.69 S/m and 4.97, 48.7, respectively.

Initially, for port-1, Figure 17a shows that the SAR is 0.0989 W/kg which is less than the value of 2 W/kg for the 10 g of tissue, as specified by IEC standards. Hence the SAR lies in the safe range, and no human tissue can be harmed. For the case of port-2 (Figure 17b), the SAR value obtained is 0.101 W/kg, which is also under the safe range, and again obeys the standard value set by the IEC (<2 W/kg). The reference power of 0.5 W has been used, and the lower value of SAR shows that the proposed antenna also qualifies for the human tissue's safety terms.

**Figure 17.** SAR of Corporate Array MIMO Placed on Belly Model: (**a**) Port-1; (**b**) Port-2.

Table 4 provides a performance comparison between the proposed MIMO design and the reported one in the literature working on the Sub-6 GHz band. It is observed that the reported work does not consider any gain improvement within the desired operating band, and apart from this, low isolation is achieved by the reported antennas. The proposed work, however, does provide a high gain antenna with a quite good total efficiency and

high isolation level within the operating band, which makes the proposed antenna quite competent for 5G-based communication applications.

**Table 4.** Comparison with reported work based on sub-6 GHz.

#### **5. Conclusions**

In this paper, a circular-shaped antenna is demonstrated with an inner circular slot, plus a rectangular slot at the right edge of it to make the proposed design resonate at the 5.57 GHz dedicated for Sub-6 GHz 5G communication applications. To achieve a desired gain, four-element array configuration is adopted which improves a bore side gain to 12.4 dB from 6.66 dB. Then, the two-port configuration is adopted such that the isolation achieved between them is more than −30 dB. The total efficiency of the proposed antenna array is observed to be more than 80% within the operating bandwidth, and the measured results of the proposed array are in good agreement with the simulated ones. Moreover, the SAR analysis is also presented for the proposed MIMO configuration; obeying the standard value of <2 W/kg for any 10 g of tissue. Thus, the proposed design becomes a potential candidate for 5G mobile phones and handheld devices. In the future, the MIMO antenna will be fabricated and tested to validate the results, and will be further extended utilizing the metamaterials concept.

**Author Contributions:** Conceptualization, S.U.; methodology, J.K.; software, J.K. and F.A.T.; validation, F.A.T. and J.K.; formal analysis, F.T.; investigation, J.K.; resources, S.U.; writing—original draft preparation, J.K. and F.A.T.; writing—review and editing, S.U., F.T. and R.R.; visualization, F.T.; supervision, S.U.; project administration, S.U.; funding acquisition, R.R. All authors have read and agreed to the published version of the manuscript.

**Funding:** This research received no external funding.

**Institutional Review Board Statement:** Not applicable.

**Informed Consent Statement:** Not applicable.

**Data Availability Statement:** The data presented in this study are available on request from the corresponding author.

**Conflicts of Interest:** The authors declare no conflict of interest.

#### **References**

## *Article* **A Compact mmWave MIMO Antenna for Future Wireless Networks**

**Muhammad Imran Khan 1,\*, Sarmadullah Khan 2, Saad Hassan Kiani 3,4, Naser Ojaroudi Parchin 5,\*, Khalid Mahmood 6, Umair Rafique <sup>7</sup> and Muhammad Mansoor Qadir <sup>8</sup>**

**Abstract:** This article presents a four-element multiple-input multiple-output (MIMO) antenna design for next-generation millimeter-wave (mmWave) communication systems. The single antenna element of the MIMO systems consists of a T-shaped and plow-shaped patch radiator designed on an ultrathin Rogers RT/Duroid 5880 substrate. The dimensions of the single antenna are 10 <sup>×</sup> 12 mm2. The MIMO system is designed by placing four elements in a polarization diversity configuration whose overall dimensions are 24 <sup>×</sup> 24 mm2. From the measured results, it is observed that the MIMO antenna provides 9.23 GHz impedance bandwidth ranging from 22.43 to 31.66 GHz. In addition, without the utilization of any decoupling network, a minimum isolation of 25 dB is achieved between adjacent MIMO elements. Furthermore, the proposed MIMO antenna system is fabricated, and it is noted that the simulated results are in good agreement with the measured results. Through the achieved results, it can be said that the proposed MIMO antenna system can be used in 5G mmWave radio frequency (RF) front-ends.

**Keywords:** MIMO; mmWave; plow-shape; polarization diversity

#### **1. Introduction**

The new communication era is based on fifth-generation (5G) technology, which has been deemed the fastest data-rate-providing technology when compared to thirdgeneration (3G) and fourth-generation (4G) technologies. The International Telecommunication Union (ITU) divided the 5G communication spectrum into two parts. One of them is named the sub-6 GHz spectrum [1], whereas the second one is called the millimeter-wave (mmWave) spectrum [2–4]. The reason for choosing sub-6 GHz spectrum is to achieve high data rates with existing communication technologies [5], but the antenna size is quite large and also the gain of the antenna will be low [6]. In contrast to this, at mmWave frequency bands, the small wavelength provides an extra advantage for designing compact antennas for 5G communication systems. In addition, the higher frequency bands provide large bandwidth, which ultimately leads to achieving a high data rate [7,8]. At mmWave spectrum, the propagation loss is high due to small wavelengths [9,10], and this problem can be solved by designing high-gain antennas. In addition, high data rates can be achieved by utilizing multiple-input multiple-output (MIMO) technology. For this purpose, many researchers have reported different MIMO antenna designs in the literature for the mmWave spectrum.

**Citation:** Khan, M.I.; Khan, S.; Kiani, S.H.; Ojaroudi Parchin, N.; Mahmood, K.; Rafique, U.; Qadir, M.M. A Compact mmWave MIMO Antenna for Future Wireless Networks. *Electronics* **2022**, *11*, 2450. https://doi.org/10.3390/ electronics11152450

Academic Editor: Christos J. Bouras

Received: 27 June 2022 Accepted: 3 August 2022 Published: 6 August 2022

**Publisher's Note:** MDPI stays neutral with regard to jurisdictional claims in published maps and institutional affiliations.

**Copyright:** © 2022 by the authors. Licensee MDPI, Basel, Switzerland. This article is an open access article distributed under the terms and conditions of the Creative Commons Attribution (CC BY) license (https:// creativecommons.org/licenses/by/ 4.0/).

In [11], a coplanar waveguide (CPW)-fed antenna array was presented for mmWave networks. The authors designed a linear MIMO array by utilizing a T-shaped radiator with a defected ground plane. The use of a defected ground plane enhanced the isolation between antenna elements, which tends to achieve high channel capacity, but it reduces the directivity of the MIMO antenna. In [12], a four-element MIMO antenna was designed for mmWave applications. A nature-inspired patch radiator and a partial ground plane with a semi-circular shell was used to achieve a wideband response. The presented results show that the single antenna provides a bandwidth of 4 GHz in the frequency range of 26–30 GHz. In the MIMO configuration, a minimum isolation of 24 dB was observed between the MIMO elements. In [13], a quad-port MIMO antenna design was presented for 5G mmWave applications. The MIMO's single element consists of a 1 × 2 array of patch radiators. To achieve wideband response and high isolation, a defected ground structure (DGS) configuration was utilized. It was observed that the use of DGS tends to achieve a wideband response from 25.5 to 29.6 GHz, whereas the isolation between antenna elements was noted to be less than 10 dB. The same kind of MIMO configuration was designed in [14]. In this design, the authors used a slotted zig-zag decoupling structure to achieve >25 dB of isolation in the band of interest. In [15], a tree-shaped patch-based MIMO antenna system was reported for future mmWave communication systems. The radiating structure was designed in such a way so that it can provide a wideband impedance bandwidth ranging from 23 to 40 GHz. It was also observed that the designed patch antenna offered a high gain of 11.45 dBi in the band of interest.

In [16], a MIMO antenna was designed by using the design technique presented in [13]. The presented MIMO antenna resonates well in the frequency range of 24.55–26.5 GHz. To achieve high isolation (>25 dB), an array of metasurface was placed above the MIMO antenna. The same kind of technique was utilized in [17]. In this design, the authors used a novel patch shape with a partial ground plane to achieve a wideband response ranging from 23.5 to 29.4 GHz. Furthermore, the isolation between the antenna elements was noted to be >20 dBi for the entire operating bandwidth. For high gain, a 2 × 2 array of metasurface was used and achieved a peak gain of 10.44 dBi. Although the MIMO structures provide high isolation and high gain, their bulky configuration restricts their use in compact communication devices. In [18], a novel MIMO antenna configuration was adopted to achieve high isolation. The MIMO design was able to operate at a very wideband frequency range of 36.83–40.0 GHz. Furthermore, an isolation of >18 dB was achieved with a peak gain of 6.5 dBi. However, the presented configuration can only be used in arbitrary-shaped communication devices. In [19], three-element and four-element MIMO antenna designs were presented for 5G mmWave applications. The authors used an inverted-L shaped patch element to achieve a wideband response in the frequency range of 26–40 GHz. In the three-element MIMO design, they employed slots in the ground plane to achieve an isolation of >15 dB between antenna elements, whereas in the fourelement design, semi-arc was designed within the slots to obtain an isolation of >22 dB. Although the presented designs offer wideband response and high isolation, they suffer due to low antenna efficiency.

Based on the above-presented discussion, in this article, a T-shaped and plow-shaped patch-based wideband four-element MIMO antenna design is presented for mmWave communication systems. The proposed design is based on a polarization diversity technique and occupies an overall size of 24 × 24 mm2, which enables its use in compact communication devices. The wideband response is achieved by utilizing a partial ground plane with a square notch. It is observed from the presented results that the MIMO antenna operates well from 22.43 to 31.66 GHz and provides measured impedance bandwidth of 9.23 GHz. Furthermore, without the use of any decoupling or isolation enhancement network, an isolation of 25 dB is achieved between adjacent antenna elements.

#### **2. Antenna Design**

#### *2.1. Single Element*

The proposed single antenna element design is shown in Figure 1. The front-side of the antenna consists of a T-shaped and plow-shaped patch radiator (see Figure 1a), whereas the back side consists of a partial ground plane with a square notch, as shown in Figure 1b. The plow-shape is comprised of a T-shaped resonator and two extended strips bent at 15◦. The partial ground plane is utilized to achieve wideband repose in the band of interest, whereas the square notch is used to enhance the impedance matching. A low-loss ultra-thin Rogers RT/Duroid 5880 dielectric substrate is used for the design of an antenna. The thickness of the dielectric substrate is chosen to be 0.254 mm, and its relative permittivity is equal to 2.2. The overall dimensions of the proposed plow-shaped antenna are noted to be 10 × 12 mm2. For the feeding of a plow-shaped patch, a 50 <sup>Ω</sup> microstrip feeding line is used, whose length and width are equal to 6.75 mm and 0.787 mm, respectively (see Figure 1a). The feed line length is extended to use the connector in an efficient manner. The rest of the design parameters with their respective values are shown in Figure 1.

**Figure 1.** Proposed single antenna element (**a**) front-view and (**b**) back-view.

(**b**)

The simulated reflection coefficient (S11) response of the proposed single antenna element is illustrated in Figure 2a. One can note from the figure that, according to −10 dB impedance bandwidth criteria, the antenna operates well from 22.24 to 31.76 GHz, providing an impedance bandwidth of 9.52 GHz and a fractional bandwidth of 35.25%. The simulated radiation and total efficiency responses of the proposed antenna are shown in Figure 2b. The radiation efficiency is noted to be >95% for the desired operating bandwidth. On the other hand, the total efficiency fluctuates in the range of 83–97%. In addition, the proposed antenna realized gain is shown in Figure 2c. It is noted that the gain of the antenna varies from 2 to 3.5 dBi in the operating bandwidth (see Figure 2c).

**Figure 2.** Simulated (**a**) S11, (**b**) radiation and total efficiency, and (**c**) realized gain of the proposed antenna.

The single antenna element was developed through a series of parametric studies. The parameters optimized to obtain the desired response are: square notch modeling in the ground plane, the angle of the open strips attached at both ends of the plow-shaped resonator, the horizontal strip of the plow-shaped resonator, and the length of the Tshaped resonator. Figure 3a shows the effect of ground slot on an antenna's performance. The parameter "Px" value has been changed from 1.3 to 1.7 mm. As the value of "Px" increases, a shift in the lower frequency is observed, as shown in Figure 3a. The optimum response is achieved at Px = 1.5 mm, where the antenna provides acceptable bandwidth and impedance matching (see Figure 3a). For values greater than 1.5 mm, the bandwidth of the antenna is going to decrease, as depicted in Figure 3a.

Figure 3b shows the parametric modeling of the side suspended strips whose length is named as "G". The performance is assessed by changing the rotation angle of the strips from 0◦ to 20◦. As seen from Figure 3b, this angle plays a vital role in producing a desired response. As the angle increases, an increase in impedance bandwidth is observed (see Figure 3b). From the presented study, the desired response is achieved at an angle of 15◦ angle, as shown in Figure 3b. The effect of parameter "D" on antenna's performance is depicted in Figure 3c. As the value of "D" changed from 2.3 to 2.7 mm, an improved impedance matching is observed on the expense of reduce antenna bandwidth. For D = 2.3 mm, the antenna offers wide impedance bandwidth but provides low impedance matching in the band of interest (see Figure 3c). The optimized response is achieved for D = 2.5 mm, as shown in Figure 3c.

Figure 3d depicts the effect of T-shaped resonator length "F" on the antenna's performance. As the length of the T-shaped resonator increases, a drastic change is observed in the impedance bandwidth, as shown in Figure 3d. For F = 1.75 mm, the antenna provides a wide impedance bandwidth, but the impedance matching is low. An acceptable performance in terms of bandwidth and impedance matching is achieved for F = 2.25 mm (see Figure 3d).

**Figure 3.** Effect of (**a**) square notch width "Px", (**b**) strip angle, (**c**) plow-shaped horizontal strip "D", and (**d**) T-shaped resonator length "F" on antenna's performance.

#### *2.2. MIMO Antenna Configuration*

The proposed single antenna element was transformed into a four-element MIMO configuration to achieve a high data rate and high channel capacity in the mmWave spectrum. Figure 4 shows the design of the proposed MIMO antenna. The MIMO antenna is designed by placing the above-presented antenna element in a polarization diversity configuration, as shown in Figure 4a, whereas the back side consists of independent partial ground planes loaded with a square notch (see Figure 4b). The dimensions of the proposed MIMO antenna are noted to be 24 × 24 mm2. From the figure, one can observe that there is no decoupling network designed between the antenna elements to achieve high isolation.

**Figure 4.** Proposed MIMO antenna (**a**) front-view and (**b**) back-view.

#### **3. Experimental Results and Discussion**

#### *3.1. S-Parameters*

To verify the simulated data, the proposed MIMO antenna was fabricated and tested using an in-house facility. Figure 5 shows the proposed antenna fabricated prototype. The Sparameters of the proposed MIMO antenna, both simulated and measured, are shown in Figure 6. For simplicity, the reflection coefficients of only two antenna elements are shown. Figure 6a shows the simulated and measured reflection coefficients of antenna-1 and

antenna-2. From the simulated reflection coefficients, shown in Figure 6a, it is observed that antenna-1 resonates from 22.2 to 31.66 GHz, whereas antenna-2 operates in the frequency range of 22.4–32.57 GHz. On the other hand, the measured results show that antenna-1 and antenna-2 provide impedance bandwidths ranging from 22.95 to 31.53 GHz and 22.43 to 31.66 GHz, respectively (see Figure 6a). The simulated and measurement isolation results are depicted in Figure 6b. The isolation between adjacent antenna elements is noted to be greater than 25 dB for the entire operating bandwidth. The discrepancies between the results may arise due to SMA connector losses, fabrication tolerances, and indoor scattering environments.

**Figure 5.** Fabricated prototype of proposed MIMO antenna.

**Figure 6.** Simulated and measured (**a**) reflection coefficients and (**b**) port isolations of the proposed MIMO antenna.

#### *3.2. Radiation Characteristics*

The far-field radiation characteristics of the presented MIMO antenna are assessed using a traditional method. As a reference antenna, a dual-ridge horn antenna with a frequency range of 15–40 GHz is employed, and the designed MIMO antenna is mounted on a turntable that is positioned on the opposite side. RF absorbers are used to cover the anechoic chamber's walls in order to eliminate reflections.

The simulated and measured gain patterns for *φ* = 0◦ and *φ* = 90◦ are shown in Figure 7. The radiation patterns are extracted at 24.5 GHz. It is observed from Figures 7a,c that antenna-1 and antenna-3 offer omnidirectional radiation properties for *φ* = 0◦ and bi-directional (monopole-like) radiation characteristics for *φ* = 90◦. The same kinds of radiation patterns are observed for antenna-2 and antenna-4 (see Figures 7b,d). In this case, the proposed MIMO antenna shows polarization diversity behavior, as shown in Figures 7b,d. In addition to this, the proposed MIMO antenna system is also able to provide pattern diversity in both the planes at the same time, which is clear from the results of Figure 7. This effect can also be observed from Figure 8 where three-dimensional (3-D) radiation characteristics of the proposed antenna are plotted. From the results of Figure 8, it can also be observed that the gain of the MIMO antenna varies in the range of 5.23–5.46 dBi.

**Figure 7.** Far-field radiation characteristics of the proposed MIMO antenna at 24.5 GHz for (**a**) antenna-1, (**b**) antenna-2, (**c**) antenna-3, and (**d**) antenna-4.

**Figure 8.** 3-D radiation patterns of the proposed MIMO antenna at 24.5 GHz for (**a**) antenna-1, (**b**) antenna-2, (**c**) antenna-3, and (**d**) antenna-4.

#### *3.3. Total Efficiency and Realized Gain*

In Figure 9a, the total efficiency of all the antenna elements is plotted, whereas the overall MIMO realized gain is depicted in Figure 9b. From Figure 9a, it can be observed that the total efficiency of the antenna elements is greater than 85% for the entire operating bandwidth. On the other hand, the simulated gain of the MIMO antenna fluctuates in the range of 5.23–5.46 dBi, whereas the measured gain varies from 4.98 to 5.66 dBi (see Figure 9b).

**Figure 9.** (**a**) Total efficiency and (**b**) realized gain of the proposed MIMO antenna.

#### *3.4. Surface Current*

The simulated surface current distribution of the proposed MIMO antenna for all the antenna elements is depicted in Figure 10. It can be noted that for all antenna elements, maximum current is distributed around the feedline, the radiating structure, and the square notch etched on the ground plane. So, it can be said that the square notch plays an important role in achieving a wideband response.

**Figure 10.** Surface current distribution for (**a**) antenna-1, (**b**) antenna-2, (**c**) antenna-3, and (**d**) antenna-4.

#### **4. MIMO Parameters**

MIMO diversity parameters are important to discuss while designing a MIMO antenna system. These MIMO parameters include envelope correlation coefficient (ECC), diversity gain (DG), and mean effective gain (MEG). ECC is the measure of how well the antennas are isolated and can be calculated using the far-field radiation characteristics of the MIMO antenna [1]. In Figure 11a, the ECC of the presented MIMO antenna system is presented. Within the desired operating band, the ECC value is observed to be less than 0.008, which follows the standard value of ≤0.5. The low value of ECC also ensures that the proposed MIMO system will have independent channel operation. Furthermore, the DG of the designed MIMO antenna is shown in Figure 11b. The DG can be calculated through ECC as mentioned in [1]. For the presented MIMO antenna system, the DG value is greater than 9.95 dB for the band of interest (see Figure 11b).

**Figure 11.** (**a**) Envelope correlation coefficient and (**b**) diversity gain of the proposed MIMO antenna.

MEG is another important MIMO parameter, which is calculated using the equation given in [3] at frequencies of 28 GHz, 28.5 GHz, and 29 GHz, respectively, for all the four antenna elements. MEG actually provides information about how much power is received by each antenna in the MIMO system. Through Table 1, the MEG of the proposed MIMO antenna system is less than −3 dB, which satisfies the convention.

**Table 1.** Mean effective gain of the proposed MIMO antenna.

The comparison among previously presented and designed MIMO antenna systems is listed in Table 2. It is noted from the table that the size of the proposed MIMO antenna is small compared to the designs of [13,14,16,19]. Furthermore, the proposed MIMO antenna system offers high bandwidth and acceptable isolation performance compared to the designs reported in [12–14,16,17].

**Table 2.** Comparison among previously presented and proposed mmWave MIMO antennas.

#### **5. Conclusions**

In this article, a wideband four-element MIMO antenna system is presented for mmWave communication systems. The MIMO antenna system is designed in such a way that it can provide both pattern and polarization diversity at the same time. From the results, it is noted that the bandwidth of the proposed MIMO antenna system is noted to be 9.23 GHz, ranging from 22.43 to 31.66 GHz. The isolation between adjacent elements and diagonal elements is observed to be >25 dB and >18.5 dB, respectively. The MIMO parameters are also evaluated and their values lie within an acceptable range. Therefore, the proposed MIMO antenna system could be used as a potential candidate for future mmWave wireless systems.

**Author Contributions:** Conceptualization, S.H.K. and N.O.P.; methodology, M.I.K. and K.M.; software, S.H.K. and K.M.; validation, N.O.P. and K.M.; formal analysis, N.O.P. and U.R.; investigation, S.K.; resources, S.K. and N.O.P.; writing—original draft preparation, S.H.K. and M.M.Q.; writing review and editing, N.O.P., U.R., and M.M.Q.; visualization, U.R. and M.M.Q.; supervision, S.H.K. and N.O.P.; project administration, K.M. and U.R.; funding acquisition, N.O.P. and U.R. All authors have read and agreed to the published version of the manuscript.

**Funding:** This project received no external funding.

**Conflicts of Interest:** The authors declare no conflict of interest.

#### **References**

## *Article* **MTM-Inspired Graphene-Based THz MIMO Antenna Configurations Using Characteristic Mode Analysis for 6G/IoT Applications**

**Sherif A. Khaleel 1,2,\*, Ehab K. I. Hamad 2,\*, Naser Ojaroudi Parchin 3,\* and Mohamed B. Saleh <sup>1</sup>**

**Abstract:** 6G wireless communications will be immersed in the future with different applications. It is expected to support all IoT services and satellite communications, and it is expected to support artificial intelligence (AI) and machine learning (ML). The THz frequency band has a vital role in 6G communication. In this study, a new graphene plasmonic two-port Terahertz (THz) MIMO antenna is analyzed by the characteristic mode theory (CMA), which gives a better insight into the physical behavior of the MIMO configurations. The proposed MIMO antenna is compact and designed on a Teflon substrate of 130 <sup>×</sup> <sup>85</sup> <sup>μ</sup>m2. The antenna provides a wide impedance bandwidth of 0.6 THz (3.2–3.8 THz). The CMA is applied to clarify the position at which the mutual coupling gives a maximum concentrated current distribution. It is mainly used to reveal the preferable MIMO antenna configuration by the usage of the model significant and model current distribution property. To reduce the mutual coupling between the radiating elements, a complementary dumbbell-structure Metamaterial (MTM) unit cell is etched in the ground plane to block the coupling mode without any affection on the dominant mode. The preferred MIMO configuration gives high isolation of −55 dB between the radiating patches. The fundamental characteristics have been discussed in detail. The proposed MIMO design offers several attractive features such as large bandwidth of 0.6 THz, low envelope correlation coefficient (ECC) of 0.000168, compact size, stable radiation, high gain of 7.23 dB, and low channel capacity loss (CCL) of 0.006. The proposed MIMO design is suitable for different applications in the THz band according to the high-performance parameters such as biomedical applications, security scanning, sensing, IoT, and 6G high-speed wireless communication systems.

**Keywords:** 6G communications; characteristic mode analysis (CMA); dumbbell-structure; graphene; metamaterial; MIMO antenna; IoT

#### **1. Introduction**

The new era of the 6 G wireless communication system has gained significant attention in recent years as a result of the growing need for greater data rates and ultra-high-speed connectivity. Around 2030, it is anticipated that the 6G wireless communication technology will be operational [1,2]. Terahertz (THz) frequency bands will be critical components of the 6G wireless communication infrastructure [3–5]. The evolution of the 6 G in terahertz frequency was the least studied band in the electromagnetic spectrum, owing to the unavailability of needed materials, sources, measurements, equipment, and detectors that operate inside the terahertz band [6]. The THz spectrum lies between the microwave and infrared in the range of 0.1–10 THz. Recently, the terahertz band has been emersed in different applications such as high data rate transmission in IoT applications [7], biomedical

**Citation:** Khaleel, S.A.; Hamad, E.K.I.; Parchin, N.O.; Saleh, M.B. MTM-Inspired Graphene-Based THz MIMO Antenna Configurations Using Characteristic Mode Analysis for 6G/IoT Applications. *Electronics* **2022**, *11*, 2152. https://doi.org/ 10.3390/electronics11142152

Academic Editors: Faisel Tubbal, Ladislau Matekovits and Raad Raad

Received: 9 June 2022 Accepted: 6 July 2022 Published: 9 July 2022

**Publisher's Note:** MDPI stays neutral with regard to jurisdictional claims in published maps and institutional affiliations.

**Copyright:** © 2022 by the authors. Licensee MDPI, Basel, Switzerland. This article is an open access article distributed under the terms and conditions of the Creative Commons Attribution (CC BY) license (https:// creativecommons.org/licenses/by/ 4.0/).

<EMAIL_ADDRESS>(N.O.P.)

applications [8,9], material characterizations [10], and high-speed secure data transmission [11]. Numerous antennas have been developed for terahertz spectrum uses, including the Leaky-Wave antenna [12], Yagi-Uda antenna [13,14], Bow-Tie antenna [15], Monopole Antenna [16], and log-periodic antenna [17]. The planar microstrip patch antenna possesses numerous advantages over the other types due to the low cost of manufacturing, low profile, simple structure, and Monolithic Microwave Integrated Circuit (MMIC) compatibility. Although, the metallic microstrip antenna suffers from low antenna performance such as gain, bandwidth, and radiation efficiency, especially on the scale of micrometer due to the low conductivity and mobility [18]. The limitation of the metallic planar microstrip patch antenna due to its drawbacks pushes researchers to the usage of 2D organic materials such as graphene, which is extracted from graphite [19–21]. The graphene utilization as a radiating patch treats the shortage of metallic materials. Graphene as a single-layer honeycomb lattice structure has excellent electrical, mechanical, thermal, and optical properties [22]. The most important property of graphene is the ability to support the propagation of the surface plasmon polariton (SPP) waves at the THz window. Additionally, graphene conductivity can be adjusted dynamically by applying an electric field via an external gate voltage, which alters the chemical potential and hence the conductivity [23]. As of today, research in millimeter-wave technology is advancing toward commercial implementation and is still spurred by the scarcity of bandwidth; the terahertz (THz) band is envisioned as the near future frontier for communication. In the present day, improved channel capacity and a high data rate are required for wireless communication to fulfill requirements. So, MIMO-communication systems operating in the THz range with data transfer rates of up to Tera-bit/sec (Tbps) are required [24]. Moreover, there are always some obstacles to the THz wireless communication frequency band such as high attenuation, Multipath fading, absorptions, and path loss. So, the MIMO system is a good candidate to face these problems.

In the MIMO antenna, the distance between elements has a very strong impact on the mutual coupling between the antenna elements. Greater mutual coupling occurs as a result of closer separation, which raises the correlation coefficients and reduces efficiency. Thus, it is required to have enough space between elements, which should be taken as λ/2, or for even better isolation it can be taken as λ/4 of the frequency to have a mutual coupling less than −15 dB [25]. Reduction in mutual coupling leads to reducing the Envelop Correlation Coefficient (ECC), increasing diversity gain (DG), and lowering the Channel Capacity Loss (CCL). It is crucial to note that graphene has inherent benefits in reducing mutual coupling effects because of its short SPP wavelength resulting from the slow propagation speed of SPP waves. As a result, graphene-based antenna arrays with the same spacing between components display fewer mutual coupling effects than metallic antenna arrays of the same distance. Different techniques and strategies are presented for the development of the decoupling methods. A neutralization line is used for isolation improvement between MIMO antenna elements, defected ground structure (DGS), electromagnetic bandgap (EBG), frequency selective surfaces (FSS), and artificial magnetic conductor (AMC) metamaterial structures that are considered effective methods to reduce the isolation between the MIMO antenna elements. The MTM structure is used as a band stop filter to reduce the coupling. Furthermore, the main parameters that the MTM used to enhance the isolation process are the structure form, the position of the unit cell, and the number of elements used. All of these parameters can be analyzed using the characteristic mode analysis (CMA) technique. CMA has become a popular technique for studying and constructing antennas, especially owing to the physical understanding acquired regarding antenna working principles without regard for any specific feeding method. Characteristic mode theory is a method of modal analysis for antennas of any form. In addition, it clearly describes the resonating frequency of certain modes, radiation patterns, and the associated mode current. Fundamental modes are extracted from an eigenvalue issue using the CMA. Calculating Z, the Method of Moments (MOM) impedance matrix, is often required to solve the following eigenvalue equation.

$$X(f\_n) = \lambda\_n R(f\_n) \tag{1}$$

where *X* and *R* are the imaginary and real parts, respectively, of the MOM impedance matrix *Z*, *J ns* are the modes or eigen-currents, and *λ<sup>n</sup>* are the eigenvalues. The eigenvalue problem is solved, for each frequency, using the procedure detailed by Harrington and Mautz [26]. Antenna analysis using the MOM approach requires an understanding of the antenna's structure and its excitation field. In the meantime, just the antenna structure is needed as a known requirement in CMA. The characteristic modes are obtained by solving the following mathematical equations:

$$
\overrightarrow{f}' = \sum\_{n} a\_n f\_n \tag{2}
$$

where *Jn* is the eigen-current and *an* is the modal weighting coefficient

$$a\_n = \frac{V\_n^i}{1 + j\lambda\_n} \tag{3}$$

where *λ<sup>n</sup>* is the eigenvalue and *V<sup>i</sup> <sup>n</sup>* is the modal excitation coefficient. The modal significance, *MSn* represented by the normalized amplitude of the current modes is calculated as follows:

$$MS\_n = \left| \frac{1}{1 + j\lambda\_n} \right| \tag{4}$$

Model significance demonstrated how each mode is near to resonance at each frequency. It reaches the maximum value of 1 at its resonance frequency (when *λ<sup>n</sup>* = 0). The modal significance ranges from 0 < *MSn* ≤ 1. The resonance mode is associated with a model significance of unity. In other words, the characteristic angle *β<sup>n</sup>* can be calculated as in Equation (5). The characteristic angle has the advantage that its value changes more quickly making it ideal for analysis.

$$
\beta\_n = 180 - \tan^{-1}(\lambda\_n) \tag{5}
$$

The CMA analysis is carried out using CST microwave studio using two different solvers, which are a surface integral solver and a multi-layer solver. The model significant (*MSn*), characteristic angle (*βn*), and the eigenvalue *λn*, all indicate the behavior of the concerned structure. At resonance mode, the value of *λ<sup>n</sup>* is equal to 0, while (*MSn*) is equal to 1 and the value of the characteristic angle (*βn*) is equal to 180◦. The inductive mode that stores the magnetic energy occurs when *λ<sup>n</sup>* > 0 or *β<sup>n</sup>* < 180◦. In contrast, the capacitive mode stores the electric energy when the value of *λ<sup>n</sup>* < 0 or *β<sup>n</sup>* > 180◦.

Herein, in this paper, a graphene plasmonic microstrip nano-antenna is proposed to be used in the 6G wireless communication system applications to operate at a 3.5 THz frequency band. A three configurations graphene plasmonic two-ports MTM microstrip patch antenna is reported based on the theory of characteristic mode analysis (CMA), which gives us the opportunity to reach the optimum design of the antenna without wasting a lot of time in the try and error iteration process. The CMA is also used to detect the optimal configuration that gives high isolation and good performance to withstand the current distribution and the model significance. In order to reduce the mutual coupling between the two-port graphene plasmonic microstrip nano-antenna, a metamaterial unit cell of Dumbbell-structure is introduced. The diversity performance is verified by the usage of the envelope correlation coefficient (ECC), channel capacity loss (CCL), and diversity gain (DG). The strategy of the proposed design is demonstrated in the following order: firstly, the design procedure of a single element graphene plasmonic nano-antenna at the resonance frequency of 3.5 THz by the usage of CMA to obtain the optimum dimensions

then a verification of the desired is carried out by another software such as HFSS and ADS to invoke the equivalent circuit model to verify the intended performance. Secondly, the graphene plasmonic two-port MIMO array antenna is used to increase the gain with three configurations. Thirdly, the characteristics of the Dumbbell-structure MTM unit cell is investigated and etched in the antenna to reduce the mutual coupling between the two elements. Finally, the diversity performance of the graphene-MIMO configuration is also carried out with and without the MTM, and this design is validated using the HFSS and ADS software.

#### **2. Graphene Patch Antenna Design**

#### *2.1. Graphene's Conductivity and Dispersion Relation*

Graphene, as a carbon-based material, has been prepared with different techniques. A single layer of graphene was extracted from the bulk graphite crystal structure using a mechanical exfoliation approach. Chemical vapor deposition (CVD), thermal exfoliation, and carbon segregation [27] are other preparation methods for the graphene layers. At the THz frequency band, this exceptional material exhibits excellent mobility, transparency, flexibility, and environmental stability. As a result, it is the best choice that can be used in antenna applications. In comparison to other nano-plasmonic materials such as gold and silver, graphene is an excellent contender for promoting the propagation of SPP waves. The dispersion relation of Transverse Magnetic (TM) of SPP depends on the conductivity of graphene given by [28].

$$-i\frac{\sigma\_s}{\omega \varepsilon\_0} = \frac{\varepsilon\_{r1} + \varepsilon\_{r2}\text{coth}\left(K\_{eff}t\_o\right)}{K\_{eff}}\tag{6}$$

where *εr*<sup>1</sup> and *εr*<sup>2</sup> are the relative permittivity of air over the graphene layer and the dielectric constant of the substrate materials, respectively. *Keff*, is the wavenumber of the *SPP* wave which can be expressed by:

$$K\_{eff} = K\_0 \left(\eta\_{eff}\right) \tag{7}$$

where *ηeff* is the effective refractive index of SPP and *K*<sup>0</sup> is the free space wavenumber *K*<sup>0</sup> = <sup>2</sup>*<sup>π</sup> <sup>λ</sup>* . According to Kubo's formula [29,30]. The graphene conductivity is the sum of both intraband and interband conductivity, and can be described as:

$$
\sigma\_{\\$} = \sigma\_{\text{inter}} + \sigma\_{\text{intra}} \tag{8}
$$

$$\sigma\_s(\omega, \mu\_c, \tau, T) = \frac{j\varepsilon^2 \left(\omega - j\tau^{-1}\right)}{\pi\hbar^2} \times \left[ \begin{array}{c} \frac{1}{\left(\omega - j\tau^{-1}\right)^2} \\\\ \int\_0^\infty \mathcal{E}\left(\frac{\partial f\_d\left(\mathcal{E}\right)}{\partial \mathcal{E}} - \frac{\partial f\_d\left(-\mathcal{E}\right)}{\partial \mathcal{E}}\right) d\mathcal{E} - \\\\ \int\_0^\infty \frac{f\_d\left(-\mathcal{E}\right) - f\_d\left(\mathcal{E}\right)}{\left(\omega - j\tau^{-1}\right)^2 - 4\left(\varepsilon/\hbar\right)^2} d\mathcal{E} \\\\ 1 \end{array} \right] \tag{9}$$

$$f\_d(\mathfrak{E}) = \frac{1}{\left[\mathfrak{e}^{(\mathfrak{E}-\mu\_{\mathbb{C}})/K\_{\mathbb{B}}T} + 1\right]} \tag{10}$$

where *T* is the temperature, *KB* is the Boltzmann constant, is the reduced Plank's constant, *μ<sup>c</sup>* is the chemical potential, *e* is the charge of electron *τ* is the relaxation time and *fd* - ࣟ is the Fermi-Dirac distribution. The first term in the preceding equation denotes graphene's intraband conductivity, whereas the second term denotes graphene's interband conductivity. Intraband conductivity dominates in the MW and THz bands, whereas interband conductivity dominates in the near-infrared and visible spectrums [31]. Graphene's intraband and interband conductivities can be expressed as:

$$\sigma\_{\rm intra}(\omega, \mu\_c, \tau, T) = -j \frac{e^2 K\_B T}{\pi \hbar^2 (\omega - j\tau^{-1})} \left[ \frac{\mu\_c}{K\_B T} + 2 \ln \left( e^{-\mu\_c/K\_B T} + 1 \right) \right] \tag{11}$$

$$\sigma\_{\rm inter}(\omega, \mu\_{\rm c}, \tau, T) = -\frac{j e^2}{4 \pi \hbar} \ln \left[ \frac{2 |\mu\_{\rm c}| - \hbar \left( \omega - j \tau^{-1} \right)}{2 |\mu\_{\rm c}| + \hbar (\omega - j \tau^{-1})} \right] \tag{12}$$

#### *2.2. Graphene Modeling*

The graphene layer is simulated and optimized using CST Microwave Studio, which is based on the Finite Element Method (FEM). This program has two built-in graphene models, namely "*graphene*" and "*graphene-Eps*". The first model uses a single layer of graphene with a thickness of 0.345 nm, whereas the second uses a multilayer of graphene sheets with varying thicknesses. In this paper, we proposed an application of a single layer of graphene sheet of 0.34 nm thickness for our presented graphene plasmonic nanoantenna. Figure 1a shows the graphical representation for the surface conductivity of the graphene layer at zero value of chemical potential and the relaxation time is 0.1 p sec at room temperature which is identical to the one presented by Llatser et al. [32].

**Figure 1.** Surface conductivity of graphene for (**a**) μc = 0 eV and τ = 0.1 ps, (**b**) different relaxation times at μc = 0 eV, and (**c**) different chemical potential values at τ = 0.1 ps T = 300 K.

The real and imaginary parts of our simulated graphene are presented in Figure 1b,c for various relaxation time (τ) and chemical potential (μc) values, respectively, using the MATLAB program. This figure demonstrates that the overall conductivity of a graphene slab is strongly reliant on the chemical potential and the relaxation time. Chemical potential is proportional to carrier density, which can be varied via an external gate voltage or chemical doping:

$$n = \frac{2}{\pi \hbar^2 v\_f^2} \int\_0^\infty \mathcal{E} [f\_d(\mathcal{E} - \mu\_c) - f\_d(\mathcal{E} + \mu\_c)] d\mathcal{E} \tag{13}$$

The relationship between the chemical potential and gate voltage is explained by [33]. Tunability is required for the wideband operation of the proposed antennas which can prevent path loss and enhance data rates in THz communication systems. Tunable antennas have been considered the best tools for manipulating the interaction of light with electrons present in materials. The fundamental capabilities of graphene antennas are vastly utilized in a broad scope of practices, including high-speed communication with high data rate, terahertz detection energy harvesting, and optical light emission.

$$V\_{\mathcal{S}} = \left[\frac{e\mu\_{\varepsilon}{}^2 h}{\pi \hbar^2 \upsilon\_f \,^2 \varepsilon\_0 \varepsilon\_r}\right] \tag{14}$$

where *h* is the substrate thickness and *ε<sup>r</sup>* is the relative permittivity of the substrate material. The conductivity of graphene can be changed with the relaxation time (τ) as follows [34].

$$
\pi \simeq \mu\_{\mathcal{S}} \hbar \sqrt{\frac{n\pi}{\varepsilon v\_f}} \tag{15}
$$

where *μ<sup>g</sup>* is the electron mobility of graphene and *vf* is the Fermi velocity. The relaxation time increases with the chemical potential values. Long relaxation time supports the propagation of the SPP wave.

#### **3. Antenna Design**

#### *3.1. Single Element Patch Antenna*

Initially, a conventional microstrip patch antenna with a graphene plasmonic layer is engraved on the top side of Teflon with a relative permittivity of 2.1, and a thickness of 10 μm. The proposed antenna is designed to resonate at 3.5 THz. The dimensions of the patch are recognized by the usage of the characteristic mode analysis (CMA) using the integral equation solver in the CST microwave studio. The optimum width and length of the patch are 32 μm and 25 μm, respectively. The single element of the patch antenna is matched to a transmission line of 50 Ω characteristic impedance using a quarter-wavelength impedance transformation. The model significance and the characteristic angel curves are shown in Figure 2a,b. Mode 1 is the resonance mode (desired mode) at a frequency of 3.5 THz in which the characteristic angle is 180◦ and the model significance in this frequency is the highest value (closely to unity). The validation process of the CST result can be achieved by the usage of HFSS and the equivalent circuit model using ADS software. The values of the electrical RLC components are calculated, and the electric circuit model of the antenna is developed using the circuit theory approach [35].

The desired current distribution specifies the optimum position for the excitation process so, the proposed structure of the single-element antenna is shown in Figure 3. The calculated value of the circuit parameters is listed in Table 1. Figure 4 shows the circuit model connection diagram of the proposed antenna. The results obtained from the HFSS and ADS circuit theory approach are in good agreement with those obtained using CST software. The |S11| curve of the graphene plasmonic antenna is illustrated in Figure 5. There is a small deviation in the |S11| curves, which may be due to the different numerical approaches used in different software. The 3D radiation pattern of the proposed graphene plasmonic antenna is depicted in Figure 6. Figure 7 shows the graphene's frequency-reconfigurable behavior of the graphene plasmonic antenna when the chemical potential applied varied between 0.5 and 2.5 eV. This figure reveals that

the plasmonic resonance frequency of the proposed antenna increases via the chemical potential and can be dynamically controlled in a wide frequency range from 2.75 to 4 THz. So, this design has dynamic reconfigurability by applying an external electric field.

**Figure 2.** (**a**) Significant and (**b**) characteristic angle of the first four modes of the proposed patch antenna.

**Figure 3.** Schematic of the graphene plasmonic patch antenna.

**Figure 4.** ADS equivalent circuit model of the proposed graphene antenna.

**Figure 5.** |S11| of the single element graphene plasmonic patch antenna with different simulation programs.

**Figure 6.** 3D radiation pattern of the proposed antenna.

**Figure 7.** |S11| parameters of the graphene plasmonic antenna for different chemical potential.

#### *3.2. Two Elements Graphene Plasmonic MIMO Antenna Configurations*

The MIMO system's advantage is that it provides data rate augmentation even in the presence of signal fading, multipath fading, and interference. The MIMO system was designed to meet the requirements of acceptable MIMO parameters because of the need to transmit more data over greater distances. Mutual coupling between elements in a MIMO array could alter the performance of a single-element antenna. The distance between the MIMO units regulates this mutual coupling, which is formed via electromagnetic interaction between them. The optimum separation distance between adjacent MIMO elements is the system's most important goal. An antenna with a large distance between elements ensures non-correlation but increases the overall antenna size, whereas a small distance will increase the mutual coupling between the elements. So, mutual coupling is an important parameter that should be studied that faces the MIMO antenna systems. The two-element graphene plasmonic microstrip MIMO patch antennas are illustrated in the following part. In order to create the optimal MIMO antenna configuration, it is necessary to first determine the orientation of the optimal elements to achieve the desired high level of isolation. Figure 8 demonstrates the proposed three configurations that are used in the antenna design. The first configuration (Ant. A) is shown in Figure 8a the two elements have a 0◦ orientation side by side with the first element. The second configuration (Ant. B) is shown in Figure 8b where the two elements are orthogonally oriented with a 90◦ between them. In the last configuration (Ant. C), the two elements have a 180◦ orientation as shown in Figure 8c. The simulated reflection coefficients |S11| and transmission coefficients |S21| versus frequency of the three MIMO configurations are investigated in Figure 9. The transmission coefficients |S21| are an important parameter while designing the MIMO system. It is the measurable parameter of the mutual coupling between the MIMO elements.

These are three possible orientations that can be used using two elements MIMO antenna. Figure 9 illustrates the |S11| parameters of the three MIMO configurations that are designed to resonate at 3.5 THz. The |S21| curves reveal that Ant. C has the lowest mutual coupling of −25.6 dB within the entire interested frequency band rather than the other configurations of Ant. B that has a −22.4 dB in addition, the Ant. A has a mutual coupling of −18 dB The three graphene patch MIMO configurations are printed on Teflon substrate with a relative permittivity (*εr*) of 2.1 and a thickness of 10 μm. The CMA is used to describe the behavior of the graphene plasmonic MIMO antenna configurations with the help of model significant (Ms), and the model current distribution for the three configurations of Ant. A, Ant. B and Ant. C.

**Figure 8.** The proposed three different MIMO antenna configurations (**a**) 0◦ orientation, (**b**) 90◦ orientation, (**c**) 180◦orientation.

**Figure 9.** S–parameters (S11 and S21) of the three different MIMO configurations.

#### *3.3. Analysis of the Graphene Plasmonic MIMO Configurations*

The evaluation of the two-element graphene plasmonic MIMO antenna structure is carried out and optimized using CMA for three different configurations. Firstly, in Ant. A structure, the model significance, and the model current distribution curves at the resonance frequency of 3.5 THz are investigated in Figures 10a and 11a, respectively. The model significance curve indicates the first four modes of operation through the entire band. It can be seen that the four modes play a significant role in the desired bandwidth (*Ms* ≥ 0.707), but modes 1 and 2 give a high contribution in the desired bandwidth rather than modes 3 and 4. Moreover, modes 1 and 2, have the highest values (close to unity). So, it is expected that one of these modes is responsible for the antenna resonation at 3.5 THz and the other one is responsible for coupling between the two-elements' graphene plasmonic MIMO antenna. The model current distribution makes the divergence between the first two modes. From the current distribution depicted in Figure 11a; mode (Jn1) is the dominant mode that is responsible for the antenna resonation and mode (Jn2) causes the coupling between the two-radiating elements of the MIMO antenna because of the

high current density between the elements which, in turn, affects the overall antenna performance negatively. Therefore, it is imperative that the characteristics of this mode need to be tuned again with a certain approach. In addition, mode (Jn3) and mode (Jn4) have no impact on the coupling between the MIMO elements. Secondly, in Ant. B, the model significant curve shown in Figure 10b illustrates the first four modes. It can be seen that all modes have the opportunity to contribute to the radiating bandwidth (*Ms* ≥ 0.707) with different levels of acceptance such as modes 1 and 3, which have the highest values of *Ms*, while modes 2 and 4, have the lowest one. So, the model significant property is not sufficient to specify all the modes and clarify which of them is responsible for the coupling, non-coupling, and resonance effect. The model current distribution in Figure 11b is a good candidate that is used to illustrate this point. This figure reveals that modes (Jn3) and, (Jn4), have a current null between the antenna elements, and mode (Jn1) is the resonance one while mode (Jn2) is the coupling mode that needs to be characterized and tuned again to reduce the mutual coupling between the graphene plasmonic MIMO antenna elements. Finally, in Ant. C configuration, the model significant curve depicted in Figure 10c demonstrates the first four modes that participate in the desired bandwidth (*Ms* ≥ 0.707). It is interesting to note that, modes 1,2, and 3 only contribute to the desired bandwidth of the antenna as they have the highest value of *Ms*. However, mode 4 can be neglected because of its low value of *Ms* within interested bandwidth. This result can be assured by introducing the model current distribution in Figure 11c. From this curve, we can overlook mode (Jn3) and (Jn4) as the current density concentration is very weak between the patches. So, mode (Jn1) is the resonance mode, and mode (Jn2) is the coupling one. The model significance and model current distribution curves of the proposed three configurations of the MIMO antenna reveal that the configuration of the designed antenna is responsible for the degree of the coupling between the two elements. The model current distribution indicates that the best configuration is Ant. C which gives a low coupling between the patches and low current density concentration in comparison with Ant. A and Ant. B.

#### *3.4. De-Coupling Structure of Dumbbell Shape MTM Unit Cell*

As the distance between the elements of the MIMO antenna is responsible for the mutual coupling issue, the performance of the antenna seriously deteriorates as this distance decreases. The three configurations of the MIMO antenna introduced above suffer from mutual coupling problems, so, a de-coupling structure is needed to overcome this problem. A complimentary new shape of Dumbbell-structure MTM unit cell is etching in the ground plane with a strong band rejection characteristic (Band-stop Filter) illustrated in Figure 12 and its dimensions listed in Table 1. The S-parameter curve depicted in Figure 13 illustrates that the Dumbbell-structure MTM unit cell has a high band-stop characteristic |S21|. The permittivity and permeability of the unit cell in Figure 14a,b reveal that this structure is an MTM unit cell with a negative value if (*ε<sup>r</sup>* and *μr*) in the desired bandwidth. The MTM unit cell of the Dumbbell-structure that is etched in the ground plane will reduce the mutual coupling between the two elements of the graphene MIMO configurations due to its band rejection characteristic at the desired frequency band. As mentioned before, in the proposed three configurations there is only one mode responsible for the coupling between the two elements (mode 2) in all configurations. So, this mode should be returned to improve the isolation. The MTM unit cell is designed to produce a current flow distribution in the opposite direction of the coupling mode current (mode 2). The characteristic mode theory (CMA) has a vital role in illustrating the behavior of the graphene MIMO systems with and without introducing the Dumbbell-structure unit cell by the usage of model significant and model current distribution. The Ant. C is the best choice to invoke the MTM unit cell on its ground plane as it has a good characteristic rather than the other configurations.

**Figure 10.** Model Significant of the proposed MIMO configurations (**a**) Ant. A, (**b**) Ant. B, and (**c**) Ant. C.

In Ant. C configuration, with the presence of an MTM unit cell as depicted in Figure 15, the model current distribution of the first four modes is shown in Figure 16. It can be seen that the mutual coupling is reduced and the path of the current through the MTM design is orthogonal to the path of the current that occurred due to the coupling mode (mode 2). So, the net interference between the two patches is remarkably reduced. This result can be assured from the model significance curve illustrated in Figure 17, we can note that the first four modes of operations and only mode 1 affects the desired bandwidth (*Ms* ≥ 0.707). Mode 2, which is responsible for the coupling is moved away from the bandwidth in the condition that the resonance mode 1 does not affect and stays the same. Moreover, modes 3 and 4 do not affect the desired bandwidth which can be seen from the model current distribution and model significant curves. At last, from the study of the CMA analysis in the three proposed configurations of graphene MIMO antenna, introducing the Dumbbell-structure MTM unit cell in the ground plane has a good impact on the MIMO antenna parameters, especially in the interference problem between the patches that enhance the overall performance of the antenna. Ant. C, in the presence of an MTM unit cell, is the preferable configuration of the two-element graphene plasmonic MIMO structure, which gives high isolation between the patches as shown in Figure 18, which presents the |S11| and |S21| of the proposed antenna with and without the presence of the MTM unit cell.

**Figure 11.** Model current distribution (Jn1–Jn4) of the proposed MIMO configurations (**a**) Ant. A, (**b**) Ant. B, and (**c**) Ant. C.

**Figure 12.** Simulation set up and dimensions for the Dumbbell-structure MTM unit cell.

**Figure 13.** S–Parameters of Dumbbell-structure MTM unit cell.

**Figure 14.** Extracted (**a**)permittivity and (**b**) permeability of Dumbbell-structure MTM unit cell.

**Figure 15.** Ant. C structure in the presence of MTM unit cell.

**Figure 16.** Model current distribution of Ant. C with the MTM unit cell.

**Figure 17.** Model significant of Ant. C with the MTM unit cell.

**Figure 18.** S–Parameters (S11 and S21) of Ant. C with and without the MTM unit cell.

The S–Parameters and the ADS for the equivalent circuit models of the proposed design are listed in Figures 19 and 20 respectively. The isolation coefficient |S21| has been reduced from −25 dB to −55 dB because of the introduction of the Dumbbell-structure MTM unit cell in the ground. The ADS software is used to validate the results by using the equivalent circuit model (ECM) analysis. The ECM consists of a two-input port with 50 Ω input impedance and is separated by the resonator circuit that represents the Dumbbellstructure MTM unit cell. The calculated values of the circuit elements are listed in Table 1. The S–Parameters of the ECM analysis is compared with the CST microwave studio. It can be seen that the ADS results are in good agreement and coincide with the CST result with a little change due to the values of the RLC components.

**Figure 19.** S–Parameters of Ant. C with CST and ADS equivalent circuit model.

**Figure 20.** Equivalent circuit model of the proposed structure of two-element MIMO configurations with the MTM unit cell.

#### *3.5. Analysis of MIMO Antenna Parameters*

In this section, we discuss the most important parameters of the MIMO antenna structure. These parameters are very crucial in the MIMO systems. As the demand for a higher data rate for long-distance transmission is one of the major motivations behind the MIMO configuration to ensure the compatibility of the graphene plasmonic MIMO antenna in the THz frequency band. The envelope correlation coefficient (ECC), diversity gain (DG), and channel capacity loss (CCL) are evaluated and discussed [36,37]. The ECC is defined as the correlation between the patches in the chosen MIMO configuration. The lower amount of ECC between the MIMO elements means higher MIMO performance. There are two methods to extract the ECC parameters, the first one is by extracting from the S–Parameters as in [38].

$$ECC = \rho\_{ij} = \frac{\left| S\_{11}^{\*} S\_{12} + S\_{21}^{\*} S\_{22} \right|^{2}}{\left( 1 - \left( \left| S\_{11} \right|^{2} + \left| S\_{21} \right|^{2} \right) \right) \left( 1 - \left( \left| S\_{22} \right|^{2} + \left| S\_{12} \right|^{2} \right) \right)} \tag{16}$$

where *S*∗ <sup>11</sup> and *S*<sup>∗</sup> <sup>21</sup> represent the conjugate of *S*<sup>11</sup> and *S*21, respectively. The second one is by extracting from the 3D radiation pattern as illustrated in [25].

$$ECC = \rho\_{ij} = \frac{|\iint\_{4\pi} \left[ F\_i(\theta, \varphi) \bullet F\_j(\theta, \varphi) \right] d\Omega|^2}{\iint\_{4\pi} |F\_i(\theta, \varphi)|^2 d\Omega \left| \iint\_{4\pi} \left| F\_j(\theta, \varphi) \right|^2 d\Omega} \tag{17}$$

where *Fn*(*θ*, *ϕ*) is the 3D field radiation pattern of the elements associated with the nth port, • denotes the Hermitian product operator and Ω represents the solid angle. The evaluation of the ECC using a far-field radiation pattern is a complex method, so the S–Parameters method is very convenient to calculate the ECC between the two antenna elements as it is faster and easy to be used as it depends only on the port parameters of the antenna system. The agreeable limit of the ECC is lower than 0.5. The proposed graphene plasmonic MIMO antenna is simulated and ECC is calculated and displayed in Figure 21a. This figure reveals the comparisons between the MIMO elements with and without introducing the complementary Dumbbell-structure MTM unit cell in the ground plane. The ECC without MTM has a value of 0.0189, while the ECC with MTM is reduced to 0.000168 through the entire bandwidth of operation. The lower value of ECC indicates a lower correlation between antenna elements. This value guarantees good MIMO performance of the proposed antenna. Another way to describe and evaluate the performance of the MIMO antenna structure is the diversity gain (DG), which describes the losses that occur in the transmitted power. The formula given in [39] is used to calculate the DG value: *DG* <sup>=</sup> <sup>10</sup>

1 − (*ECC*)

<sup>2</sup> (18)

**Figure 21.** (**a**) Envelop correlation coefficient (ECC), (**b**) Diversity gain (DG), and (**c**) channel capacity loss (CCL) of Ant. C with and without the presence of the MTM unit cell.

It can be seen that the DG depends on the value of ECC, moreover, equation (18) states that the small value of ECC produces a higher value of DG and makes the MIMO antenna suitable for diverse applications. Figure 21b depicts the DG value of the proposed MIMO structure in the presence and absence of the MTM structure it has a value of 9.999 and 9.904, respectively.

Channel capacity loss (CCL) is one of the most significant MIMO parameters that describe the potential loss of channel capacity due to the correlation between the MIMO antenna elements, in other words, CCL defines the maximum limit of information transmission rate without significant loss. It should be less than 0.5 bits/s/Hz for a well-designed MIMO configuration. The CCL is indicated in (19) as reported in [39].

$$\text{CCL} = -\log\_2[\det(\Phi\_k)],\tag{19}$$

$$
\Phi\_k = \begin{bmatrix} \rho\_{11} & \rho\_{12} \\ \rho\_{21} & \rho\_{22} \end{bmatrix} \tag{20}
$$

$$\rho\_{\bar{i}\bar{i}} = \left(1 - \left(\left|S\_{\bar{i}\bar{i}}\right|^2 + \left|S\_{\bar{i}\bar{j}}\right|^2\right)\right), \quad and \tag{21}$$

$$\rho\_{i\bar{j}} = -\left(S\_{i\bar{i}}^{\*}S\_{i\bar{j}} + S\_{i\bar{j}}^{\*}S\_{j\bar{j}}\right)\_{\prime} \tag{22}$$

where Φ*<sup>k</sup>* indicates the correlation matrix at the receiving antenna. Figure 21c depicts the simulated CCL of the proposed MIMO antenna. It recorded the specified limit for the proposed MIMO configuration. A comparison between the CCL curves with and without introducing the MTM unit cell. So, it has a value of 0.006 and 0.840, respectively. It can be confirmed that the proposed design of graphene plasmonic MIMO configuration delivers a higher transmission data rate in any scattering environment. Figure 22 illustrates the representation of the far-field radiation pattern component of the E and H plane at 3.5 THz. The E and H plane radiation patterns are represented in terms their components. The proposed graphene plasmonic MIMO antenna radiation patterns are illustrated in the presence and the absence of the MTM unit cell. The proposed MIMO configuration exhibits a little change in the E and H planes as shown in Figure 22. At last, the E and H patterns indicate that no significant difference between the main lobe pattern with and without the Dumbbell-structure MTM unit cell.

**Figure 22.** Far-field radiation of Ant. C with and without MTM unit cell (**a**) E-plane and (**b**) H-Plane.

#### *3.6. Performance Analysis with State-of-the-Art of Printed THz Antennas*

In this part, comparisons between proposed optimized Ant. C design with previously reported antennas has been performed for the terahertz frequency band. The comparison is conducted based on performance parameters such as gain, size of the antenna elements, and resonance frequency. Moreover, the MIMO parameters such as ECC, DG, and CCL are tabulated in Table 2. Based on the presented data, we can conclude that the proposed graphene plasmonic MIMO antenna has the best isolation and ECC. Although the proposed design presents a high gain with all compared references, the design in [40] has a higher gain, but gives low isolation and CCL. Finally, the proposed antenna demonstrates a superior MIMO performance to that of the other listed designs in Table 2. So, it is a good candidate to be used in 6G for mobile generation and medical applications. It has a suitable compact size to easily integrate with on-chip integrated circuits.

**Table 2.** Comparison of the proposed antenna performance with the state-of-the-art works.

#### **4. Conclusions**

In this paper, a high-performance graphene plasmonic two-port MIMO antenna has been introduced to be used in terahertz wireless communication applications. In the theory of characteristic mode analysis (CMA) with the help of a complementary Dumbbellstructure MTM unit cell, the mutual coupling between the antenna elements has been significantly reduced. The MTM structure acts as a band stop filter in the desired bandwidth. It is etched in the ground plane so as to reduce the mutual coupling from −25 dB to −55 dB in the presence of the MTM unit cell. The CMA has been used to study the behavior of the antenna modes of operations and clarify the mode responsible for the coupling between the elements and detect the optimum position of this mode which can be illuminated by the MTM structure without any distortion in the main antenna resonance mode (dominant mode). Furthermore, the MIMO antenna parameters such as ECC, DG coefficients, and CCL are investigated and presented to confirm the suitability of the proposed graphene MIMO antenna for different THz applications. The proposed MIMO antenna resonates at 3.5 THz with a return loss of −35 dB and exhibits a wide impedance characteristic bandwidth covering 3.2 to 3.8 THz frequency spectrum in the THz regime (BW = 600 GHz). The designed MIMO antenna maintains maximum isolation of −55 dB at the resonance frequency of 3.5 THz. The good MIMO performance is ensured by achieving the MIMO performance parameters (ECC = 0.000168, DG = 9.999, and CCL = 0.006) within acceptable limits. The proposed MIMO antenna design in the future can be used as a massive MIMO and used the reconfigurable intelligent surfaces to be suitable for the 6G communication technology The proposed MIMO antenna is suitable for high-speed short-distance indoor communication applications such as video-rate imaging, biomedical imaging, sensing, and security scanning in the terahertz frequency band.

**Author Contributions:** S.A.K. raised the idea, wrote the manuscript, and perform the numerical simulations, E.K.I.H. and N.O.P., revised the manuscript and perform the theoretical analysis, N.O.P. and M.B.S., discussed the numerical results in the main manuscript; S.A.K. and E.K.I.H. contribute to the simulation results and the validation, improved the revised version. All authors have read and agreed to the published version of the manuscript.

**Funding:** This research received no external funding.

**Institutional Review Board Statement:** Not applicable.

**Informed Consent Statement:** Not applicable.

**Data Availability Statement:** All the data have been included in the study.

**Conflicts of Interest:** The authors declare no conflict of interest.

#### **References**

## *Article* **Ultra-Wideband Pentagonal Fractal Antenna with Stable Radiation Characteristics for Microwave Imaging Applications**

**Muhammad Abbas Khan 1, Umair Rafique 2,\*, Hüseyin ¸Serif Savci 3, Anis Nurashikin Nordin 4, Saad Hassan Kiani 5,6 and Syed Muzahir Abbas 7,\***

**Abstract:** For microwave imaging applications, a design for an ultra-wideband (UWB) fractal antenna is presented. The antenna design is composed of a pentagonal fractal patch radiator fed by a modified co-planar waveguide (CPW) ground plane. It is built on a low-loss Rogers RT/Duroid 5880 dielectric substrate with a dimensions of 24 <sup>×</sup> <sup>30</sup> <sup>×</sup> 0.787 mm3. According to the measurements, the designed antenna offers a fractional bandwidth of 123.56% ranging from 3 GHz to 12.7 GHz. In addition, a maximum gain of 3.6 dBi is achieved at 8.5 GHz. From the results, it is also observed that the proposed antenna structure attains constant radiation characteristics in the operating bandwidth, which is useful for microwave imaging applications. The time domain analysis of the proposed design is also performed, and it is observed that the designed antenna offers a group delay of ≤1.5 ns, which ensures minimum pulse distortion.

**Keywords:** microwave imaging; ultra-wideband; fractal antenna; co-planar waveguide; fractional bandwidth; group delay

#### **1. Introduction**

Near-field microwave imaging has gotten a lot of attention in recent years, especially for detecting and localizing cancer tissues in the human body [1]. Microwave imaging offers promising solutions for a number of biological applications [2] when compared to existing approaches such as ultrasound [3] and mammography [4]. In this technique, short pulses of low power are emitted by antennas towards the human body. The backscattered radiation is then collected by the antennas and processed to form an image. One of the examples is shown in Figure 1. From the figure, one can observe that multiple antennas operating in the near field are placed around the breast. The antennas forming the array are sequentially selected to transmit pulses into the breast. Then, the backscattered signals are collected by the receiving antennas. Finally, a suitable signal processing technique is applied to backscattered signals for image reconstruction.

Microwave imaging antennas should have a wide bandwidth and be able to provide constant radiation performance. Furthermore, the antenna's time domain performance

**Citation:** Khan, M.A.; Rafique, U.; Savci, H. ¸S.; Nordin, A.N.; Kiani, S.H.; Abbas, S.M. Ultra-Wideband Pentagonal Fractal Antenna with Stable Radiation Characteristics for Microwave Imaging Applications. *Electronics* **2022**, *11*, 2061. https:// doi.org/10.3390/electronics11132061

Academic Editor: Andrea Randazzo

Received: 9 June 2022 Accepted: 29 June 2022 Published: 30 June 2022

**Publisher's Note:** MDPI stays neutral with regard to jurisdictional claims in published maps and institutional affiliations.

**Copyright:** © 2022 by the authors. Licensee MDPI, Basel, Switzerland. This article is an open access article distributed under the terms and conditions of the Creative Commons Attribution (CC BY) license (https:// creativecommons.org/licenses/by/ 4.0/).

should be sufficient to allow transmitted pulses to pass through human tissues with minimal signal distortion. Ultra-wideband (UWB) technology may be useful in meeting these criteria as the pulse duration in UWB systems is often on the scale of a few nanoseconds, resulting in a bandwidth of a few gigahertz. In addition, the antenna size should be small enough so that it can easily be integrated into microwave imaging systems. To accommodate this requirement, researchers presented a variety of antenna designs. From the presented designs, planar antennas received a lot of attention due to their small size and light weight.

**Figure 1.** A typical near-field microwave imaging system for breast cancer detection [5] (Reprinted from an open access article published under Creative Commons Attribution 4.0 License).

For UWB microwave imaging systems, the authors in [6] presented a modified square monopole antenna design. In the ground plane, a pair of loop sleeves was designed, which tend to achieve an impedance bandwidth in the frequency range of 3.19–11.03 GHz. Although the designed antenna offers a wide impedance bandwidth, its efficiency is decreasing with an increase in frequency. For body-centric imaging applications, Danjuma et al. [7] designed a co-planar waveguide (CPW)-fed quasi-cross slot-based circular monopole antenna. The presented antenna can operate at frequencies ranging from 3 GHz to more than 11 GHz. A Hibiscus petal-like patch antenna was presented in [8] for microwave imaging systems. To obtain an impedance bandwidth in the frequency range of 3.04–11 GHz, a tapered microstrip feed line and a partial trapezoid ground plane were used. The performance of the antenna was evaluated in the time domain and a high value of group delay was observed for the side-by-side configuration, which is not useful for microwave imaging applications. In [9], an octagonal planar antenna was designed to detect malignant tissues in the human body. To achieve UWB response, the radiating element was fed by a 50 Ω microstrip feeding line and a step impedance transformer. For improved radiation characteristics, a circular slot including a symmetrical metal cross was designed in the octagonal radiator. From the presented configuration, the authors achieved an impedance bandwidth of 12 GHz from 3 GHz to 15 GHz.

Lee et al. [10] introduced a UWB-resistive dipole antenna for microwave imaging applications. They demonstrated that the designed dipole antenna offers minimal internal reflections compared to other UWB antenna structures. They employed chip resistors on both arms of the dipole antenna according to the technique presented in [11], and observed a UWB response ranging from 1 GHz to 10 GHz. However, this kind of configuration led to a large antenna size and was complex in nature. A square fractal monopole antenna with a slotted partial ground plane was presented in [12] for imaging purposes. The designed antenna-measured results demonstrate that it offers a fractional bandwidth (FBW) of 117.88% from 3.1 GHz to 12 GHz. A modified square monopole antenna design was presented in [13] for medical applications. With the use of modified square patch and inverted U-shaped parasitic elements, the authors achieved a 5.2 GHz impedance bandwidth from 3.8 GHz to 9 GHz. Another square monopole antenna with a T-slot-loaded ground plane was designed in [14] for UWB microwave imaging applications. In addition to the T-slot, a parasitic element with a *π* shape was placed beneath the microstrip line to achieve a UWB response in the frequency range of 2.91–14.72 GHz.

Ojaroudi and Ghadimi [15] utilized the same technique as presented in [14] to design a multi-resonance UWB antenna for microwave imaging applications. In this design, an E-shaped slot was etched in the ground plane, and an E-shaped parasitic structure was designed behind the microstrip feeding line. The presented configuration extends the upper-frequency limit of the antenna from 10.3 GHz to 15 GHz. The designs presented in [14,15] were able to provide a UWB response, but their radiation characteristics were not stable over the operating bandwidth. In [16], three T-shaped slots were etched in the ground plane to achieve resonance ranging from 2.96 GHz to 15.8 GHz. Lasemi and Atlasbaf [17] designed compact UWB fractal antennas for breast tumor detection. The Sierpinski gasket configuration was used for one antenna, and a ring-inscribed hexagonal fractal antenna was used for the other. According to the measurements, the designed antennas operate well from 2.95 GHz to 12 GHz. However, the time domain performance of the presented designs reveals that the transmission phase of the antennas was not stable over the operating bandwidth, which ultimately led to pulse distortion.

A negative index mematerial-based planar antenna was designed in [18] for microwave imaging applications. To achieve a compact size, the edges of the triangular patch were loaded with four left-handed metamaterial unit cells. To achieve negative permittivity and permeability, each metamaterial unit cell was composed of a modified split-ring resonator (SRR) and a capacitance-loaded strip. Measurement results show that the presented antenna operates well from 3.4 GHz to 12.5 GHz and offers a maximum gain of 5.16 dBi at 10.15 GHz. The same technique was utilized in [19] to design a compact planar metamaterial antenna. In this design, a trapezoidal patch radiator was used, which was fed through a stepped tapered microstrip feed line. The presented antenna was able to operate in the frequency range of 4.23–14 GHz. In [20], a wearable textile antenna with a full ground plane was designed for breast cancer detection. Photonic bandgap (PBG) and substrate-integrated waveguide (SIW) technology were utilized for antenna design. The antenna was fed using a grounded CPW (GCPW) through a SIW transition. The results demonstrate that the antenna offered an impedance bandwidth of 21 GHz from 7 GHz to 28 GHz. The presented antenna operates at higher frequencies that are not useful for biomedical applications. In addition, the use of PBG and SIW technologies offers complexities in the fabrication process.

 The results demonstrate that the antenna ranges from 3 GHz to 12.7 GHz, with a peak gain of 3.6 dBi at 8.5 GHz. Furthermore, from the time domain analysis, it is noted that a group delay for both face-to-face and side-by-side configurations is ≤1.5 ns, ensuring minimum pulse distortion, which is a fundamental criterion of near-field microwave imaging systems. To further demonstrate the suitability of the proposed antenna, a comparison is presented among proposed and previously published UWB antennas meant for microwave imaging applications (see Table 1).

From Table 1, it can be noted that all the antennas have fractional bandwidth >100%. The designs presented in [9,10,20] have maximum and approximately equal fractional bandwidth, but their electrical dimensions are large in comparison to the proposed design. The rest of the antennas exhibit low fractional bandwidth compared to the proposed antenna design. Furthermore, the proposed fractal antenna offers a high impedance bandwidth compared to the designs presented in [6–8,10,12,17,18].

**Table 1.** Comparative analysis among proposed and previously presented planar monopole antennas for microwave imaging applications.

#### **2. Pentagonal Fractal Antenna Design**

Figure 2 illustrates the geometry of the proposed UWB antenna. The proposed design is composed of a modified CPW-fed pentagonal fractal radiator, printed on a low-loss Rogers RT/Duroid 5880 dielectric substrate with a relative permittivity (*εr*) of 2.2. The thickness of the dielectric substrate for the antenna design is chosen to be 0.787 mm. The overall dimensions of the antenna are 24 × 30 mm2. From Figure 2, one can note that a trapezoidal ground plane is used for better impedance matching in the band of interest. Furthermore, the CPW feed technique is utilized because it is easy to design and fabricate and it provides low losses [21]. The rest of the design parameters' values are as follows: *WS* = 24 mm, *LS* = 30 mm, *Wf* = 2.5 mm, *Lf* = 12 mm, *t* = 1.5 mm, *g* = 0.3 mm, *Lg* = 5.5 mm, *Wg* = 4.2 mm, *Rb* = 3 mm, *Re* = 1 mm, *S*<sup>1</sup> = 9.6 mm, *S*<sup>2</sup> = 7.6 mm, *S*<sup>3</sup> = 5.4 mm, *S*<sup>4</sup> = 3 mm, *R*<sup>1</sup> = 7 mm, *R*<sup>2</sup> = 5 mm, and *R*<sup>3</sup> = 2.8 mm.

**Figure 2.** Geometry of the proposed CPW-fed pentagonal fractal antenna.

The design steps of the proposed CPW-fed pentagonal fractal antenna are shown in Figure 3a–d. In the first step, a CPW-fed circle-inscribed pentagonal radiator is designed, as shown in Figure 3a. The pentagonal radiator's dimensions are calculated using the following expression [22].

$$f\_l = \frac{7.2}{(L+r+p)\times k} \quad \text{GHz} \tag{1}$$

where

$$L = 2A, \quad r = \frac{A}{4} \tag{2}$$

where *A* denotes the radius of the of the pentagonal radiator, which is equal to 9.6 mm. The parameters *fl* and *k* in Equation (1) correspond to the lowest operating frequency of the antenna and the effective relative permittivity (*εref f*) of the dielectric substrate, respectively.

**Figure 3.** Design evolution of the proposed CPW-fed pentagonal fractal antenna (**a**) Step 1, (**b**) Step 2, (**c**) Step 3, and (**d**) Step 4 (proposed).

The antenna design shown in Figure 3a is simulated in the CST Microwave Studio and the respective reflection coefficient (*S*11) is given in Figure 4. From the figure, it is observed that the design shown in Figure 3a provides a dual-band response at 4 GHz and around 12 GHz. In the second step, another circle-inscribed pentagonal patch is added in the Step-1 design, as shown in Figure 3b. This modification tends to excite higher order modes and ultimately leads to a UWB response in the band of interest (see Figure 4, Step-2). For improved matching, another pentagonal radiator is added in the Step-2 design, as depicted in Figure 3c. This design offers improved impedance matching in the band of interest, as shown in Figure 4. To further optimize the response, another pentagonal patch is designed with the Step-3 design, as shown in Figure 3d. In this step, the ground plane shape is changed to a trapezoidal shape (see Figure 3d). This modification led to a UWB response in the frequency range of 3.38–12.89 GHz, as shown in Figure 4.

**Figure 4.** Reflection coefficient (*S*11) of different antenna designs shown in Figure 3.

The simulated surface current distribution plots for the proposed antenna are shown in Figure 5. The surface current is plotted for three different frequencies, i.e., 4 GHz, 8 GHz, and 12 GHz. It is evident from the figure that at 4 GHz, the dense current is distributed on the feed line, on the edges of the ground plane, and in the lower portion of the patch radiator, as shown in Figure 5a. For higher frequencies, the current is uniformly distributed on the surface of the radiator and modified the CPW ground plane (see Figure 5b,c).

**Figure 5.** Simulated surface current distribution of the proposed CPW-fed pentagonal fractal antenna at (**a**) 4 GHz, (**b**) 8 GHz, and (**c**) 12 GHz.

A parametric study was performed to understand the behavior of the proposed UWB antenna by varying the gap between the microstrip feeding line and the ground plane, denoted as *g*, and the gap between the ground plane and the main radiator, denoted as *t*. From Figure 6a, it is observed that when the value of *g* is increased from 0.3 mm to 0.9 mm, the bandwidth of the antenna decreases. The maximum impedance matching is obtained for *g* = 0.3 mm (see Figure 6a). On the other hand, when the value of *t* is changed from 0.5 mm to 1.5 mm, the impedance bandwidth of the antenna improves, as shown in Figure 6b. For *t* >1.5 mm, the bandwidth of the antenna is reduced (see Figure 6b).

**Figure 6.** Effect of parameter (**a**) *g* and (**b**) *t* on the antenna's performance.

#### **3. Fabricated Prototype and Measurements**

To verify the simulation results, the designed antenna was fabricated using the miling process. The prototype of the fabricated antenna is shown in Figure 7a. The *S*<sup>11</sup> of the proposed antenna was measured by Rohde Schwarz (R&S) using the vector network analyzer (VNA) ZVA-67. The simulated and measured *S*<sup>11</sup> characteristics are shown in Figure 7b. The antenna has a computed impedance bandwidth of 9.51 GHz in the frequency range of 3.38–12.89 GHz and a measured impedance bandwidth of 9.7 GHz in the frequency range of 3–12.7 GHz (see Figure 7b). Some discrepancies are observed between the simulated and measured results, which could arise due to fabrication tolerances, SMA connector losses, and the scattering measurement environment.

**Figure 7.** (**a**) Fabricated prototype. (**b**) Simulated and measured *S*<sup>11</sup> of the proposed CPW-fed pentagonal fractal antenna.

The simulated and measured radiation characteristics of the proposed fractal antenna are shown in Figure 8. For both the YZ and XZ planes, radiation characteristics are observed at three different frequencies: 4 GHz, 8 GHz, and 12 GHz. From the figure, it is observed that the designed antenna exhibits bi-directional radiation properties for the YZ plane and omnidirectional characteristics for the XZ-plane. It is also observed from Figure 8 that the radiation patterns are almost constant for the entire operating bandwidth, which is a useful property for microwave imaging applications.

**Figure 8.** Simulated and measured radiation characteristics of the proposed CPW-fed pentagonal fractal antenna at (**a**) 4 GHz, (**b**) 8 GHz, and (**c**) 12 GHz.

Figure 9 shows the realized gain and efficiency of the proposed antenna. The peak gain of the antenna is noted to be 3.6 dBi at 8.5 GHz, while the minimum gain is 0.6 dBi at 3 GHz, as shown in Figure 9. Furthermore, the efficiency of the antenna is noted to be >80% for the entire operating bandwidth (see Figure 9).

**Figure 9.** Realized gain and efficiency of the proposed CPW-fed pentagonal fractal antenna.

#### **4. Time Domain Analysis**

For near-field microwave imaging applications, it is necessary to assess the performance of the antenna in the time domain [23]. For this purpose, two identical antennas are placed 30 cm apart from each other in two different configurations, i.e., face to face and side by side, respectively, as shown in Figure 10a,b, in which both the antennas work in the transceiver mode.

To excite the antennas, a Gaussian pulse, centered at 8 GHz with a frequency band of 1–15 GHz, is utilized. Figure 11a shows the normalized amplitudes of both input and output signals for both configurations. The cross-correlation between transmitted and received signals is calculated using Equation (3), which is known as the fidelity factor (FF) [24].

$$FF = \max\left[\frac{\int\_{-\infty}^{\infty} \chi\_t(t)\chi\_r(t+\tau)d\tau}{\int\_{-\infty}^{\infty} |\chi\_t(t)|^2 dt \int\_{-\infty}^{\infty} |\chi\_r(t)|^2 dt}\right] \tag{3}$$

where *Yt*(*t*) and *Yr*(*t*) represent transmitted and received signals, respectively; and *τ* represents the group delay. The FF values for face-to-face and side-by-side configurations are noted to be 96.85% and 88.52%, respectively. The high value of FF for the face-to-face configuration shows that the transmitted signal will be less distorted.

The group delay for both the configurations, shown in Figure 10, is presented in Figure 11b. The group delay is defined as *the negative rate of change of the transfer function phase, φ*(*ω*) *with respect to frequency*. The group delay quantifies the transition time taken by the signal to travel through a device [23]. Mathematically, it can be calculated as:

$$\tau(\omega) = -\frac{d\phi(\omega)}{d\omega} = -\frac{d\phi(\omega)}{2\pi df} \tag{4}$$