**Lightning Modeling and Its Effects on Electric Infrastructures**

Editors

**Massimo Brignone Daniele Mestriner**

MDPI • Basel • Beijing • Wuhan • Barcelona • Belgrade • Manchester • Tokyo • Cluj • Tianjin

*Editors* Massimo Brignone Naval and Electrical Engineering Department, University of Genoa Italy

Daniele Mestriner Naval and Electrical Engineering Department, University of Genoa Italy

*Editorial Office* MDPI St. Alban-Anlage 66 4052 Basel, Switzerland

This is a reprint of articles from the Special Issue published online in the open access journal *Applied Sciences* (ISSN 2076-3417) (available at: https://www.mdpi.com/journal/applsci/special issues/Lightning Modeling Effects Electric Infrastructures).

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**Abstract**

STEM education employs an interdisciplinary pedagogical approach that merges the subjects of science, technology, engineering, and mathematics into traditional disciplines covered across all levels within primary and secondary school curricula. STEM topics span the United Nation's Sustainable Development Goals (SDGs) and naturally weave STEM disciplines throughout, focusing on the importance of developing a global STEM-literate workforce capable of responding to the worldwide challenges presented today and into the future. STEM-focused frameworks provide students with the knowledge and skills they need to be successful in the twenty first century and beyond, preparing them for the rapidly changing technological and scientific landscapes. This chapter explores STEM as a meta-discipline, documenting the origins and emergence of STEM education in the United States, provides definitions of STEM literacy and includes primary and secondary implementation models.

**Keywords:** science, technology, engineering, mathematics, STEM, STEM education, interdisciplinary frameworks, interdisciplinarity, twenty first century skills, primary STEM pedagogy, secondary STEM pedagogy, sustainable development goals (SDGs)


**Novel Step-Down DC–DC Converters Based on the Inductor–Diode and Inductor–Capacitor–Diode Structures in a Two-Stage Buck Converter**

**Mauricio Dalla Vecchia 1,2,\*, Giel Van den Broeck 1,2, Simon Ravyts 1,2 and Johan Driesen 1,2**

Received: 14 January 2019; Accepted: 19 March 2019; Published: 22 March 2019

**Abstract:** This paper explores and presents the application of the Inductor–Diode and Inductor-Capacitor-Diode structures in a DC–DC step-down configuration for systems that require voltage adjustments. DC micro/picogrids are becoming more popular nowadays and the study of power electronics converters to supply the load demand in different voltage levels is required. Multiple strategies to step-down voltages are proposed based on different approaches, e.g., high-frequency transformer and voltage multiplier/divider cells. The key question that motivates the research is the investigation of the aforementioned Inductor–Diode and Inductor–Capacitor–Diode, current multiplier/divider cells, in a step-down application. The two-stage buck converter is used as a study case to achieve the output voltage required. To extend the intermediate voltage level flexibility in the two-stage buck converter, a second switch was implemented replacing a diode, which gives an extra degree-of-freedom for the topology. Based on this modification, three regions of operation are theoretically defined, depending on the operational duty cycles δ2 and δ1 of switches S2 and S1. The intermediate and output voltage levels are defined based on the choice of the region of operation and are mapped herein, summarizing the possible voltage levels achieved by each configuration. The paper presents the theoretical analysis, simulation, implementation and experimental validation of a converter with the following specifications; 48 V/12 V input-to-output voltage, different intermediate voltage levels, 100 W power rating, and switching frequency of 300 kHz. Comparisons between mathematical, simulation, and experimental results are made with the objective of validating the statements herein introduced.

**Keywords:** DC–DC converter; experimental verification; Inductor–Diode; Inductor–Capacitor–Diode; nonisolated; step-down; two-stage buck converter; voltage regulation

#### **1. Introduction**

In the last century, AC systems predominated as the most widespread method for energy generation, transmission, and distribution. For regular AC generation, large power plants, e.g., coal, hydro, and nuclear, have been used to centrally generate energy. Alternatives, such as wind and solar generation, are more frequently being installed to decentralize the energy production and lower the CO2 emitted by the energy production [1,2].

Wind farms (AC generation) are more common in remote regions (on and off-shore) where the wind is conducive for generation. Solar (DC generation), on the other hand, started to become viable for application in high populated areas where the sun irradiation is high enough to achieve maximum power generation. DC systems started to be developed and implemented in pilot projects and real

applications because they have proved to be more efficient than AC systems in terms of transformation steps [3], costs [4], and power transfer capability [5]. Applications of DC system such as in microand nanogrids [6,7], datacenters [8,9], and aircrafts, where redundancy and backup is mandatory, are currently on the rise. DC system technology can also be implemented in isolated/remote/emerging regions [10]. In these systems, redundancy normally is indispensable in order to keep the system operational even if a component and/or a converter fails. Furthermore, DC loads are common especially in low voltage/low power applications, for instance, in computer and battery chargers, USB and USB-C ports, and LED lights. These type of loads are commonly found in offices and residences.

Discussions among the voltage levels that are implemented in these systems are extensively presented in the literature [11], for instance, with a voltage between 350 and 400 V for distribution in unipolar or bipolar [6,12,13] DC configurations to supply high power equipments. The 48 V level is [14,15] normally adopted as an intermediate bus stage to supply the low voltage/low power loads aforementioned. The 48 V level is also getting more common in battery systems for Electric Vehicles (EV), and it is considered for residential applications due to its user safety approach.

A DC picogrid, based on state-of-the-art architectures [3,6,16], is depicted in Figure 1, which emulates a building and/or residence with a DC grid implementation. In this grid, a bipolar DC network is realized (P0N poles), which has the possibility of injecting the energy generated by the panels in the AC grid or, alternatively, directly supply DC loads bypassing the DC–AC–DC transformation.

A lot of effort has been spent to propose new topologies and systems to interconnect PV generation in AC or DC grids with, respectively, microinverters [17–19] and microconverters (DC power optimizers) [20–24] which, in essence, are step-up converters since the PV panel output voltage is lower than the AC grid or DC bus voltages. The control aspects of this system and MPPT strategies are common subjects in the literature [20,25–27].

Different strategies to step-up the voltage have been studied and proposed in the literature, such as: isolated step-up topologies [17,21,28], coupled-inductor, nonisolated topologies [29–35], nonisolated topologies with Capacitor–Diode/switch structure [19,36–38], nonisolated topologies with Inductor–Diode/switch [39,40] and nonisolated topologies with Capacitor–Diode/switch and Inductor–Diode/switch structures [41,42]. These structures are known, respectively, as switched-capacitors and switched-inductors. An adaptation of the Inductor–Diode structure is proposed in [43], where a capacitor replaces one diode and changes the gain of the structure for step-up applications. Each of the above-mentioned strategies to step-up voltages has its advantages and disadvantages. For instance, isolated topologies and coupled inductor topologies have the flexibility of, based on the turns ratio of the magnetic device, select the suitable output voltage requirement, although it is known that these magnetic components are, together with heatsinks, the bulkiest components in a converter. Switched capacitor and switched inductors do not bring this flexibility once the gain is defined based on the operational duty cycle of the active switches. Inductors are more reliable components than capacitors, giving an advantage in terms of reliability, while capacitors are more compact and more suited for high power density applications.

In step-down conversion ratio converters, the Capacitor–Diode/switch structures are commonly used to improve the step-down ratio of the converters [44–46]. On the other hand, the implementation of an Inductor–Diode/switch and Inductor–Capacitor–Diode structures in the step-down conversion ratio is not extensively discussed in the literature, with few exceptions that explore both step-up and step-down structures [39]. The structures evaluated in this paper are shown in Figure 2. The Inductor–Diode structure shown in Figure 2 can offer a redundant path if one diode of the structure fails in open-circuit, as proposed in [47]. In steady-state, the inductors can be considered as current sources and, due to the structure operability discussed in Section 4, mitigate the implementation of current sensors for control purposes, since the current balance between the inductors is naturally achieved without any control strategy (switched-inductor characteristic). The only sensor that might be consider for implementation is a voltage sensor, if the application requires a regulated and constant

output voltage level for the entire load range of operation. For instance, a contraction technique was proposed to control a regular step-down converter [48].

With these considerations, the objective of this paper is to investigate the influence of the Inductor–Diode (ID) and Inductor–Capacitor–Diode (ICD) structures in the step-down conversion stage from 48 V to 12 V input-to-output voltage. The main contributions of this paper are (1) an analysis of the influence of the ID and ICD structures in a step-down conversion system; (2) the presentation of a full theoretical analysis of the converters generated; (3) applying the ID and ICD structures in a quadratic buck converter; (4) analyzing the intermediate voltage level behavior for different operational points, showing the intermediate voltage flexibility of the circuit, which is beneficial for a voltage regulation point of view; (5) creating a flexible test bench to be able to validate the proposed converters and; (6) experimentally validatation with waveforms of the topologies studied in the paper.

The paper is organized as follows; Section 2 presents the analysis of the ID and ICD structure in a regular single-stage buck converter and its impact on the gain of the converter. Section 3 presents the two-stage buck converter, known as quadratic buck converter [49], to enable the converter applying the ID and ICD structures to achieve the required output voltage levels. The operational principle of the topologies generated by the proposed intra-exchangeability is explained in Section 4. Section 5 explores the simulation results of the proposed topologies, showing the intermediate and output voltage levels that are obtained for each possible configuration. Section 6 describes the experimental test bench built for further validation, showing experimental waveforms that validate the theoretical analysis presented in Sections 3 and 4. Section 7 presents the main conclusions and future work suggestions.

#### **2. Impact of Inductor–Diode and Inductor–Capacitor–Diode Structures in a Step-Down Converter**

This section has the objective of investigating the influence of the ID and ICD structure in a step-down buck converter. Inductor I, shown in Figure 3, is replaced by ID and ICD structures and the PWM modes are described herein. The main difference between the ID and ICD structures is that the capacitor of the ICD is also charged during the magnetization of the inductors. When discharging, the capacitor also discharges, providing energy to the load. The capacitor stores part of the energy that is delivered to the load, which helps decrease the size of the inductors, since less energy is stored in their magnetic field. Additionally, the capacitor changes the voltage applied across the inductors during demagnetization, which changes the gain of the structures, as described herein.

**Figure 1.** A basic DC picogrid, highlighting the PV generation and DC step-up conversion; the bipolar DC network (P0N); the battery energy storage system (BESS); and the DC loads supplied by DC–DC step-down converters.

**Figure 2.** (**a**) Inductor–Diode (ID) structure and its charge and discharge modes and (**b**) Inductor–Capacitor–Diode (ICD) structure and its charge and discharge modes.

**Figure 3.** DC–DC Buck Converter with Point-of-Connection 1 (PoC1) to interchange inductor (I), Inductor–Diode (ID) and Inductor–Capacitor–Diode (ICD) structures.

#### *2.1. PWM Mode 1*

In this PWM mode, the switch S1 commutates from 'open' to 'close', changing the equivalent circuit of the converter. The equivalent circuit for this mode is shown in Figure 4a,b for the ID and ICD applications in PoC1, respectively. During this mode, the voltage applied in the inductor terminals VL1 and VL2 is positive, due to the step-down mechanism, and energy is magnetically stored in the inductors during the process. The voltage applied across the inductors terminals has the same level (internally in ID and ICD structures), which leads to similar current level in the inductors of the structures. In the ID structure, the inductors store energy during this PWM Mode while in the ICD structure the inductors and the capacitor of the structure store energy.

#### *2.2. PWM Mode 2*

In this PWM mode, the switch S1 commutates from 'close' to 'open'. The equivalent circuit for this mode is shown in Figure 4c,d for the ID and ICD structures, respectively. During this PWM mode, the voltage applied across the inductors terminals VL1 and VL2 is negative, since diode D1 is forward biased, leading to a demagnetization of these inductors. The current across both inductors is equal as these components are in series, which guarantees the same negative voltage level across the inductors of the structures during the demagnetization. The demagnetization mechanism for both structures is similar, since in both structures the inductors are connected in series and the same current flows through the passive components.

*Energies* **2019**, *12*, 1131

From the above PWM mode analysis, the theoretical voltage profile for all the inductors in both structures are presented in (1). These equations are applied in (2) to obtain the conversion gain of the structures, in steady-state.

$$\begin{cases} \begin{array}{c} \text{ID} \\ V\_{L1} = V\_{L2} = V\_{IN} - V\_{\text{OUT}} \text{ during } \delta\_1 \\ V\_{L1} = V\_{L2} = -\frac{V\_{\text{OUT}}}{2} \text{ during } (1 - \delta\_1) \end{array} \end{cases} \quad \begin{cases} \text{ID} \\ V\_{L1} = V\_{L2} = V\_{\text{C1}} = V\_{\text{IN}} - V\_{\text{OUT}} \text{ during } \delta\_1 \\ V\_{L1} = V\_{L2} = \frac{V\_{\text{IN}}}{2} - V\_{\text{OUT}} \text{ during } (1 - \delta\_1) \end{cases} \quad \text{(1)}$$

$$
\overline{V\_L} = \int\_0^{T\_S} v\_L dt = 0 \tag{2}
$$

#### *2.3. Theoretical Gain in Continuous Conduction Mode*

To obtain the theoretical voltage gain profile of the topologies, operation in steady-state is considered, which allows the application of (1) in (2) to obtain the ideal gain of the structures. After some mathematical manipulation, the theoretical expression that represents the gain of the Inductor–Diode (ID) and Inductor–Capacitor–Diode (ICD) structures, respectively, is graphically represented in Figure 5. From Figure 5, it is noticed that the operational voltage range tackled in this paper cannot be realized with the ICD structure, since its gain range is, ideally, between [0.5 ; 1]. To overcome this, the quadratic buck converter—a nonlinear step-down converter generated by two buck converters connected in series with each other—as depicted in Figure 6, is the base topology used to generate the converters proposed in this paper. This converter presents an intermediate stage, giving an extra degree of freedom for implementation, since two voltage levels are generated in the same converter and both can be used to supply loads.

Only one switch is implemented on the power stage of the conventional quadratic buck, which has the benefit of control strategy simplification; and the intermediate voltage level is in the function of the duty cycle applied to switch S1. If a second switch is introduced in the power stage, the degree of freedom for voltage regulation in the intermediate stage is guaranteed by the duty cycle of the second switch S2, although the control strategy might increase in complexity, if applicable. Since the objective of the topologies is to improve the flexibility of the conversion ratio and the overall system, the quadratic buck with two switches is the converter that will be extensively explored in the next section, together with the intra-exchangeable PoC structures presented in Figure 3.

#### **3. Double-Switch Two-Stage Quadratic Buck Converter for Voltage Regulation**

As previously mentioned, the two-stage buck converter, most known as quadratic buck converter, is the base topology used for the intra-exchangeable methodology used herein for testing multiple configurations, based on the ID and ICD structures, applied for voltage regulation. Since a second stage is now implemented in the power circuit, the intra-exchangeability increases the amount of topologies that can be generated. More specifically, with three structures and two stages, the number of topologies generated is 32 = 9, where only the conventional quadratic buck is already explored in the literature. Figure 6 exemplifies the possible configurations that can be obtained from the proposed intra-exchangeability between a regular inductor (I) and the ID and ICD structures. When an I structure is connected to PoC1 and PoC2, the regular quadratic buck is obtained. In total, nine topologies are generated from this method and, since two switches are implemented in the power stage with independent duty cycles, three regions of operation can be explored and will be briefly discussed herein. The regions of operation are highlighted in Figure 6, named as region 1, when duty cycle δ2 is higher than duty cycle δ1; region 2, when δ1 is higher than δ2; and region 3, when both duty cycles are equal. The switch realization is based on a MOSFET with an antiparallel diode, as shown in Figure 6.

**Figure 4.** (**a**) PWM mode 1 for ID structure; (**b**) PWM mode 1 for ICD structure; (**c**) PWM mode 2 for ID structure, and (**d**) PWM mode 2 for ICD structure.

**Figure 5.** Continuous conduction mode graphical gain representation.

#### *3.1. Region 1 (δ2 > δ1)*

Operating in Region 1, the two stages of the quadratic buck structure operate independently, leading to intermediate and output voltages, depending on the duty cycles δ2 and δ1. The intermediate voltage across capacitor Cinter depends on the structure connected in PoC2 and the duty cycle δ2 to command switch S2. The output voltage, on the other hand, depends on both structures connected in PoC2 and PoC1, since the intermediate voltage level is the input voltage of the first stage. The equations presented in Figure 5 are reused to find the intermediate and output voltages generated by this integration. Table 1 summarize the analysis, showing the theoretical gain profile of the intermediate stage and output stage in function of the input voltage of the circuit and the duty cycle of switches S1 and S2. For instance, if the ID structure is applied in PoC1 and the I structure in PoC2, the intermediate and output voltages of the circuit are, respectively, related to equations *δ*2 and *δ*2 2*δ*1 1+*δ*1 , , which means that the intermediate voltage is regulated by the nominal set point of the duty cycle applied in S2 and the output voltage depends on the duty cycles applied to S1 and S2. The analysis considers that the duty cycles are independent, and with no phase-shift between each other.

#### *3.2. Region 2 (δ2 < δ1)*

Operating in this region, the second stage of the quadratic buck depends on the first stage, since the duty cycle of switch S1 is higher than the duty cycle of switch S2. Consequently, the intermediate capacitor continues to discharge even if the switch S2 is in 'off'-state (due to the antiparallel diode between drain-source points of S2). In this operational region, the duty cycle δ1 defines the intermediate voltage level and the output voltage level. Table 2 summarizes the voltage levels in this region of operation.

#### *3.3. Region 3 (δ2 = δ1 = δ)*

Region 3 is the region when both switches operate with the same duty cycle (δ2 = δ1 = δ) and the duty cycles are in phase with each other. For this case, the intermediate and output voltages are subject to the operational duty cycle of the circuit. The theoretical analysis, shown in Section 4, is related to operation in Region 3. Table 3 summarize the voltage gain of the topologies generated by this intra-exchangeability.

**Figure 6.** Quadratic Buck with two Point-of-Conncetions (PoCs) implemented and possible configurations achieved by the intra-exchangeability between I, ID, and ICD structures. Additionally, the operational regions for different duty cycle possibilities and switch realization.

**Table 1.** Theoretical Gain Analysis for all possible combinations between I, ID, and ICD structures in the quadratic buck configuration operating in Region 1. Intermediate and output voltage profiles in function of the input voltage of the circuit.

where *δ*2 and *δ*1 are the duty cycle of switches S2 and S1, respectively.

**Table 2.** Theoretical Gain Analysis for all possible combinations between I, ID, and ICD structures in the quadratic buck configuration operating in Region 2. Intermediate and output voltage profiles in function of the input voltage of the circuit.

**Table 3.** Theoretical Gain Analysis for all possible combinations between I, ID, and ICD structures in the quadratic buck configuration operating in Region 3. Intermediate and output voltage profiles in function of the input voltage of the circuit.

Although nine different topologies are realized in this analysis, only six different output voltage profiles are obtained, since the topologies I-ID/ID-I, I-ICD/ICD-I and ID-ICD/ICD-ID provide the same output voltage profile, while different intermediate voltage profiles are observed. Figure 7 shows the graphical representation of the output voltage profiles presented in Table 3. It is noticed that, due to the equations of the voltage profile shown in Figure 5, the highly nonlinear behavior of the quadratic buck converter is attenuated when the ID or ICD structures are used in one of the PoCs. The gain of the topologies, mathematically expressed in Table 3 and graphically in Figure 7, is less nonlinear if compared with the conventional quadratic converter, which is interesting for control purposes. These converters, however, are not suitable for applications that require high step-down capability, since the ID and ICD structures decrease the step-down capability of the structure, as shown in Figure 5 for application in a regular buck and Figure 7 in the quadratic version. Although the component part count increases for ID and ICD implementation, it also increases the flexibility to achieve multiple intermediate voltage levels to supply different loads.

On comparison of Table 2, related to region 2, and Table 3, related to region 3, it is noticed that the gain of the structures are the same. As mentioned in the previous subsection, this behavior occurs because the antiparallel diode of switch S2 continue to conduct even if switch S2 is in 'off'-state until switch S1 commutates from 'close' to 'open'. This analysis concludes that operational regions 2 and 3 generate similar PWM modes and, consequently, the same intermediate and output voltage levels while region 1 is distinguished by the independence between the stages. Section 4 will further discuss the operational principle of the double switch quadratic buck with ID and ICD structures, for operation in Region 3.

#### **4. Operational Principle of the Double Switch Quadratic Buck Converter in Region 3**

Similar to Section 2, where the one-stage I, ID, and ICD topologies have been introduced, this section introduces the PWM modes of the two-stage I, ID, and ICD topologies, where the PWM signal remains the same. The phase shift between δ2 and δ1 is considered zero (in phase with each other) for the specific analysis but if differs from zero, the gain behavior shown in Tables 1–3 will differ, since the phase-shift will influence the PWM stages and, consequently, the gain of the structures. The equivalent circuit for the first and second stages, considering all the possibilities between the exchangeable circuits, are present in Figure 8a,b, respectively. In order to evaluate the main characteristics of the topologies, the equivalent circuits presented in Figure 8 are used to mathematically evaluate the voltage and current stresses in all power components. This information is crucial to further dimension the components for practical implementation.

The voltage stress across each power semiconductor are extracted from the PWM mode stages in Figure 8 and summarized in Table 4. The voltage stress across switch S1 and diode D1 is affected by the structure applied in PoC2, for instance, if the I structure is applied to PoC2, the voltage stress across switch S1 is defined by (1 + *δ*)*VIN*, while if the ID structure is applied in PoC2, the voltage stress across switch S1 is defined by equation (1+3*δ*) (1+*δ*) *VIN*. Similarly, the voltage stress across diodes D3/D5/D7, connected to PoC1, are related to the structure connected in PoC1. For instance, maintaining PoC2

fixed with the ID structure, if the ID structure is connected to PoC1, the voltage across the diodes D3/D5 is expressed by *δ*2 (1+*δ*) 2 *VIN* and, if PoC1 is changed for an ICD structure, the voltage across diodes D3/D5 is expressed by *δ* (1+*δ*)*VIN*. Switch S2 and diode D2 stresses are independent on the structures and are equal to the input voltage VIN. The voltage stress across diodes D3-D4-D5-D6-D7-D8, which are the diodes implemented in the ID and ICD structures, are related to the operational duty cycle, input voltage, and in which PoC the structures are implemented.

Similarly, the PWM modes also provide the RMS current stress in each power semiconductor. To evaluate the RMS current stress in all components, equation (3) is introduced, where ix1 is the current that the semiconductor conducts during the first PWM mode and ix2 is the current conducted during the second PWM mode. Applying (3), the current stress in each semiconductor is characterized and presented in Table 5. Different than the voltage stress, the current stress of switches S2 and D2 depend on the structure connected to PoC2 but are independent of the structure connected to PoC1. For instance, the RMS current stress across switch S2 is expressed by *IIN*(1−*δ*) *δ* ,1 *δ* if the I structure is connected to PoC2, and expressed by *IIN*(1−*δ*) 2*δ* ,1 *δ* if the ID structure is connected. It is also noticed that the current is independent on PoC1, in this case.

The analysis abovementioned is important in order to select the components for experimental validation, since the RMS current and voltage stress are the most important characteristics to select MOSFETs and diodes. The next section presents a numerical simulation for the proposed converter performed in PLECS (4.0.2, Plexim GmbH, Zurich, Switzerland).

$$i\_{xRMS} = \sqrt{\frac{1}{T\_S} \left( \int\_0^{DT\_S} i\_{\mathbf{x}1}^2 dt + \int\_{DT\_S} i\_{\mathbf{x}2}^2 dt \right)} \tag{3}$$

**Figure 7.** Continuous conduction mode graphical gain representation.

**Figure 8.** (**a**) Equivalent circuit for PoC 2 and PoC 1 applying I, ID, and ICD structures during the first PWM mode and (**b**) equivalent circuit for PoC 2 and PoC 1 applying I, ID, and ICD structures during the second PWM mode.

**Table 4.** Voltage stress sum up for all power semiconductors in the power stage. Equations are in function of the input voltage.

**Table 5.** RMS Current stress sum up for all power semiconductors in the power stage. Equations are in function of the input current.

#### **5. Numerical Simulations**

A numerical simulation was carried out to show the available intermediate voltage levels when an output voltage of 12 V is set as standard output voltage. The 12 V voltage level can supply low voltage/low power loads to, for instance, LEDs or to charge small batteries (unidirectional power flow to charge, since the topologies are unidirectional); this is one of the main reasons to fix the output voltage on this level. The simulation and experimental parameters are presented in Table 6.

Different intermediate voltage levels are generated, depending on the structure connected in PoC2 and the operational duty cycle of the converter, as shown in Table 3. The simulation model was develop in PLECS. The only configuration that is not able to be simulated and experimentally validated in this paper is the ICD-ICD structure, since the input/output voltage level is 48 V/12 V and the theoretical gain of this structure is 0.25 < ICD-ICD < 1, making it impossible to provide the required output voltage.

A simulation is performed for each configuration and the results are shown in Figure 9. The conventional quadratic buck with I-I is show in Figure 9a, generating an intermediate voltage level of 24 V. According to Table 3, the I-ID and ID-I configurations generates the same output voltage but different intermediate voltage levels for operation with the same duty cycle. This behavior is proved via simulation and presented in Figure 9b,d, where the voltage level of 20.25 V (I-ID) and 28.5 V (ID-I) are obtained for a duty cycle of 42.2%.

Similarly, the topologies I-ICD and ICD-I operate with the same duty cycle (36.6%) and input/output voltages, however the intermediate voltage level changes since is applied different structures in PoC2. Simulation results shown in Figure 9c,g confirm the analysis, since an intermediate voltage of 17.6 V (I-ICD) and 32.8 V (ICD-I) are obtained and available to supply loads in different voltage ranges in a flexible way. The structures ID-ICD and ICD-ID are also evaluate via simulation for similar input/output voltages. Intermediate voltages of 19.2 V (ID-ICD) and 30 V (ICD-ID) are obtained and shown in Figure 9f,h, respectively, validating the theoretical analysis presented in Table 3. Additionally, the last configurations present similar overall gain (δ) with the regular buck converter, however they present an additional voltage level that can be used to supply additional loads in different voltage levels. The drawback of the structure, compared with the conventional buck, is the component part count, since the number of inductors and diodes in the power stage increase.

Finally, the ID–ID structure is briefly discussed herein. Figure 9e shows the voltage levels obtained from simulation. Similarly with the conventional I-I quadratic buck, an intermediate voltage level of 24 V is obtained, however an operational duty cycle of 33.3% is used to match the desired output voltage of 12 V.

The simulations were performed with a switching frequency of 300 kHz (switching period of 3.33 μs), since this frequency was used to calculate the I, ID, and ICD inductances for further implementation of the structures. All the inductances were select to present a 20% ripple current limit across the inductors in nominal power. A summary of the theoretical inductor current expressions in each scenario is present in Table 7, in function of the input current IIN. From the PWM mode stages discussed in the previous section and shown in Figure 8a,b, the inductors of the ID and ICD structures charge its terminals in parallel and discharge in series, which guarantees natural balance between the internal currents of the structure. This characteristic enable the non-use of current sensors to control the inductor current, since this control is guarantee directly by the ID and ICD structures, also known as switched-inductor structures. The drawback, as previously mentioned, is the increase in the number of components in the power stage.

With the objective of validating the equations shown in Table 7, a comparison between the theoretical analysis and numerical simulation results was performed and shown in Table 8. The input/output voltage of the circuit is 48 V/12 V and a rated input power of 100 W. The different configurations operate with their respective duty cycles to generate an output voltage of 12 V (check Figure 7 for duty cycle). The comparison shows that the theoretical equations match the simulation results, which proves the validity of the equations to select the correct inductances for experimental verification.

**Table 6.** Simulated and experimental parameters for validation.

**Table 8.** Inductor current comparison: theoretical/simulation.

#### **6. Experimental Verification and Test Bench**

#### *6.1. Region 2 (δ2 < δ1)/Region 3 (δ2 = δ1 = δ)*

This subsection will discuss the results obtained for operation in Region 3 (δ2 = δ1 = δ). The topologies, discussed in Sections 3 and 4, were realized for further validation. A test bench was built for experimental verification of the configurations discussed herein. Figure 10 shows the test bench setup, together with a simplified bench schematic. It is shown in Figure 10 that a function generator (equipment (4)) is used to generate the PWM signals to drive S1 and S2. The experimental results shown herein are in open-loop operation, without a control strategy to regulate the intermediate and output voltage levels. Figure 11 shows the converter realization (middle) surrounded by the PoCs (PoC2 and PoC1) developed for each configuration (the inductors were selected based on the 20% maximal current ripple for each structure).

The PoCs, developed to test the different configurations, were designed separately following the plug-n-play methodology. As shown in the converter realization in Figure 11, the PoC2 and PoC1 are actual points of connection, where different external boards are connected to test different inductor configurations (I, ID, and/or ICD). Points 1 and 2, for PoC2, and 3 and 4, for PoC1, are available to connect the external PoCs. Additionally, Table 6 present the parameters defined for the experimental setup, in which the switching frequency of 300 kHz was select to simulate and experimental validation.

When comparing the waveforms presented in Figure 9a–h (simulation) and in Figure 12a–h (experimental), it can be seen that the voltage levels in all topologies are similar in shape (DC shape) and in amplitude. Figures 9 and 12 present the input, intermediate, and output voltages of the converter for the following configurations (PoC2-PoC1); (a) I-I, (b) I-ID, (c) I-ICD, (d) ID-I, (e) ID–ID, (f) ID-ICD, (g) ICD-I, and (h) ICD-ID. Small deviations are observed between the amplitude values in simulation and in the experimental waveforms due to the nonidealities inherent in the components selected to build the prototype. We also noticed that spikes of voltage appears in all voltage levels. This behavior is common in circuits that operate in high frequency, since internal parasitics of the components and also the parasitic inductances related to tracks and soldering points start to become prominent. Additionally, the commutation between 'on'- and 'off'-stages of switches S1 and S2 also generate internal oscillations in the circuit.

**Figure 9.** Simulation results showing the different intermediate voltage levels obtained by exchanging the structures for an input/output voltage of 48 V/12 V. PoC2-PoC1: (**a**) I-I; (**b**) I-ID; (**c**) I-ICD; (**d**) ID-I; (**e**) ID–ID; (**f**) ID-ICD; (**g**) ICD-I; and (**h**) ICD-ID, respectively.

**Figure 10.** Test bench for experimental realization with (1) input source, (2) output load, (3) quadratic PoC converter, (4) PWM generator, (5) voltage measurements, and (6) scope.

**Figure 11.** Converter realization (middle) surrounding by PoCs (PoC2 and PoC1) developed to experimental validation of the proposed topologies.

**Figure 12.** Voltage levels for different structures configurations. From top to bottom and left to right: (**a**) I-I with δ = 0.5; (**b**) I-ID with δ = 0.422; (**c**) I-ICD with δ = 0.366; (**d**) ID-I with δ = 0.422; (**e**) ID-ID with δ = 0.333; (**f**) ID-ICD with δ = 0.25; (**g**) ICD-I with δ = 0.366 and; and (**h**) ICD-ID with δ = 0.25.

As mentioned in Section 3, the gain analysis performed for the operation in region 2 and 3 results in the same gain expressions. The experimental results shown herein are for operation in region 3, but can be extended for the case when region 2 is in operation.

As discussed previously, is was shown that for different operating points and configurations connected in PoC2 and PoC1, different intermediate voltage levels are obtained for an input/output voltage of 48 V/12 V. This voltage, which varies from 33 V to 18 V depending the configuration, is available in the intermediate stage and can be used to supply loads in a broad range. Depending the required intermediate voltage, the best suited configuration can be selected to connect in PoC2 and PoC1. The ICD structure, as shown in Figure 5, limits the gain in the region between [0.5 ; 1], and can be used for a small voltage regulations with a certain accuracy. The ID structure, on the other hand, presents a nonlinear gain behavior in the range between [0 ; 1]. For the same duty cycle, this structure can be used to achieve gains in between the regular buck with I structure and the buck with ICD structure, with the disadvantage of a nonlinear gain.

#### *6.2. Region 1 (δ2 > δ1)*

For operation in region 1, an extra PWM mode is performed between the normal PWM modes 1 and 2, during the switching period. This extra PWM mode guarantee the independence between stages 1 and 2, leading to the gain presented in Table 1 for all possible PoC configurations. We noticed that the intermediate gain depends only on the duty cycle δ2, while the output gain is relate to duty cycles of both stages, since the converters are cascade connected.

Figure 13 shows the PWM modes for operation in region 1 for the I-ICD configuration. Nevertheless, this analysis can be extended for all configurations. As previously mentioned, an extra PWM mode arises during the switching period (PWM mode 2 in this analysis). Switch S1—the component which is subject to the greatest voltage stress among the power components—as highlighted in Table 4, blocks only the intermediate voltage level during the switch transient between 'on' and 'off' while when operates in regions 2 and 3, the switch blocks the sum between the intermediate voltage and the input voltage. This phenomenon enables the reduction of switching losses in this component once the voltage that the component blocks during the switching transient has a lower value compared with the normal operation, although the maximum voltage that the switch needs to block remains in the same level as shown in Table 4.

Figure 14 presents an experimental verification of the stages discussed previously and shown in Figure 13. Figure 14a presents a waveform with the significant voltage levels of the circuit. As expected, an intermediate voltage of 19.5 V is achieved for a duty cycle of 40% applying the I structure in PoC2. Additionally, Figure 14a also presents the second voltage regulation stage made by the ICD structure connected in PoC1, operating with a duty cycle of 25%, in which a voltage of 11.2 V is achieved at the output of the converter. Figure 14b presents the PWMs applied to the circuit and its switching period, together with the input/output voltage of the circuit. As expected, input and output voltages are in accordance with the theoretical analysis above presented.

Finally, Figure 14c shows the experimental results of the voltage stress across the active semiconductors S1 and S2. For the same duty cycle abovementioned, the voltage stress across switch S2 is independent of the operational duty cycle and equal to the input voltage, as shown in Figure 14c. Additionally, the stress across switch S1 was also evaluated and presented in Figure 14c. It is verified that, as theoretically explained, the voltage that the switch blocks during transition between 'on' and 'off' stages is the intermediate voltage across Cinter, although the maximum voltage stress is related to the sum of input and intermediate voltages (verify PWM mode 3 between points 2–3 in Figure 13). Moreover, the switching losses generated by the overlap between current and voltage during the transition time decrease due to the lower blocking voltage during this transition. For operation in region 1, the addition of switch S2 with independent duty cycle improves the flexibility of the system and decreases the switching losses across switch S1.

**Figure 13.** PWM modes with the equivalent circuits during one switching period when δ2 > δ1.

**Figure 14.** Experimental results for operation in region 1 for operation with δ2 = 0.4 and δ1 = 0.25 and switching frequency of 300 kHz: (**a**) input, intermediate and output voltages; (**b**) input and output voltages together with the respective duty cycles; and (**c**) PWMs and voltage stress across switches S1 and S2.

As discussed previously, the introduction of the ID and ICD structures in a step-down configuration (one-stage buck or quadratic buck) enable the possibility to change the voltage level of these structures. When applied in a one-stage buck converter, the ICD structure present similar voltage gain -1+*D* 2 as the converter proposed in [50]. The structures, however, present different functionalities, since the abovementioned structure is a switched capacitor based converter with one switch, three diodes, one inductor, and three capacitors in the power stage while the structure, herein presented, is a switched inductor based topology with one switch, three diodes, two inductors, and one capacitor in the power stage. In terms of component part count number, the one-stage ICD structure present lower amount of components compared with the switched capacitor version that present similar gain [50]. The ID structure, on the other hand, presents a nonlinear step-down behavior with one extra diode and one less capacitor in the power stage (compared with the above-mentioned converters).

The ID structure is also explored in step-up configurations [39] and, when applied in a step-down structure, it changes the structure to a one composed by two inductors and two diodes, which leads to a gain of *D* 1+*D* , performing a higher step-down ratio them the ID structure. A topology with different switched capacitor/inductor structure is proposed for similar voltage gains [44] Comparing these topologies with the quadratic-buck configuration applying the ID and I structures (see Table 3), the gain of these structures is 2*D*∗*D* 1+*D* . The proposed structures, although implement more components in the power stage, present a higher step-down ratio for duty cycles lower than 0.5. For duty cycles above 0.5, the proposed converter present lower step-down capability with higher component part count.

The one-stage buck converter present similar gain (D) compared with the quadratic buck applying ID and ICD structures in PoC1 or PoC2 (see Table 3). For a component part count point of view, the conventional buck converter present more benefits compared to the structures above-mentioned, since the conventional buck converter present the lowest part count structure for a DC–DC step-down configuration. On the other hand, the converters proposed herein present multiple output ports, e.g., intermediate and output voltages, and, according to the analysis presented herein, if a second switch S2 is implemented, these voltage levels can be independently regulated (operation in Region 1), which is beneficial for an application point of view.

The major advantage that the topologies proposed herein have is the flexibility to adjust the intermediate voltage level according with the project specifications. The experimental results presented in Figures 12 and 14 show that, for the same input/output voltage, different intermediate voltage levels can be obtained depending on the operational regions (Region 1, 2, or 3) and the structure applied to PoC1 and PoC2. The main disadvantage, however, is the number of components in the power stage. When applied in a quadratic buck converters, the ID and ICD structures increase the number of semiconductor and magnetics in the power stage. This feature is undesired in some applications that power density is required, since extra space is required to place all the components. As previously mentioned, the ID and ICD structures are suited for lower voltage regulations and are not applicable if high step-down conversion ratios are required.

#### **7. Conclusions**

The analysis in this paper shows the influence of the ID and ICD structures, previously applied in step-up topologies, in a step-down configuration from 48 V to 12 V input-to-output voltage. To achieve the required voltage levels, a quadratic buck structure was implemented since the first analysis showed that the ID and ICD structures have a lower voltage regulation capability compared with the conventional I structure and, if a single stage buck converter is considered for application, the ICD structure cannot achieve the required output voltage level.

A complete theoretical analysis with PWM stages for all possible regions of operation, voltage gain, voltage/current profiles, and simulation is shown for all possible configurations generated by the substitution of the structures in PoC2 and PoC1. Voltage and current stress were evaluated for dimensioning and component selection to subsequently build the prototype for experimental validation. The analysis showed that the ID and ICD structures change the voltage and current profile depending on which PoC is connected. It is proven that the addition of a second switch S2, together with the operation in Region 1, decreases the voltage stress during turn-off of switch S1. The main advantage of the proposed approach is that the intermediate voltage level can be selected according to the application, since switch S2 is implemented and the ID and ICD structures provide different voltage profiles that can be desired for specific operational duty cycles.

The main advantages of the proposed topologies compared to the conventional buck and quadratic buck converters are as follows.

A test bench was assembled and experimental results showing that the ID and ICD structures can be used for small voltage adjustments in a step-down application were presented. Simulation and experimental waveforms were presented to validate the theoretical analysis in Section 4. The ID and ICD proved to be effective structures for applications where small voltage adjustments are required, but did not prove being useful solutions when a high step-down voltage level is required due to its gain characteristics.

#### **8. Patent**

A patent related to this work was filled in on the 15th of May 2018, with application number GB1807795.8 at the UKIPO (reference [47]).

**Author Contributions:** M.D.V.: Conceptualization, Investigation, Methodology, Validation, and Writing—Original Draft. G.V.d.B and S.R.: Validation and Revision. J.D.: Supervisor, Resources, and Revision.

**Funding:** M.D.V. and S.R. were partially funded by the EFRO SALK Solsthore Project. G.V.d.B was funded by a Ph.D. grant of Research Foundation Flanders (FWO).

**Conflicts of Interest:** The authors declare no conflicts of interest.

#### **References**

© 2019 by the authors. Licensee MDPI, Basel, Switzerland. This article is an open access article distributed under the terms and conditions of the Creative Commons Attribution (CC BY) license (http://creativecommons.org/licenses/by/4.0/).


*2.2. Instrumentation*

All the fluorescence measurements were recorded using a spectrofluorometer (Shimadzu, RF-6000, Kyoto, Japan). BPA adsorption data were recorded on a UV–Vis spectrophotometer (Shimadzu, Kyoto, Japan). The magnetization measurements of the FeOx MNPs, MNP/QDs and MNP/QD@MIPs were carried out using a vibrating sample magnetometer (VSM, Lake Shore 7307, Columbus, OH, USA) at room temperature. The morphology of the QDs was characterized by a JEM-2100F high resolution transmission electron microscopy (TEM, JEOL, Tokyo, Japan). The shape and structures of the magnetic nanomaterials were examined using a H-800 transmission electronic microscopy (Hitachi, Tokyo, Japan). The Fourier transform infrared (FT-IR) spectra were determined on a Vertex 70v spectrometer (Bruker, Karlsruhe, Germany).

#### *2.3. Synthesis of Amino-Modified ZnS: Mn2*+ *QDs and Carboxyl-Functionalized MNPs*

The MEA-modified Mn2+-doped ZnS QDs (ZnS: Mn2+ QDs@MEA) and the carboxylfunctionalized FeOx magnetic nanoparticles (FeOx@COOH MNPs) were synthesized based on previously reported methods with some modification [45–47]. The preparation procedures are presented in the Supplementary Materials.

#### *2.4. Synthesis of FeOx*/*ZnS Nanoparticles*

The FeOX/ZnS NPs was synthesized via an EDC/NHS reaction process. Briefly, 20 mg of FeOx@COOH MNPs were first dispersed in 20 mL citrate bu ffer solution (0.02 mol L−1, pH = 6.4) to prepare magnetic fluids. Then, 2 mL of magnetic fluids were added in 50 mL EDC/NHS activating agent (1:1.2 g L−1). The above mixture was stirred for 2 h, followed by 30 mg of as-prepared amino-modified QDs being added and incubated at 30 ◦C for 20 h. The brown solutions were collected by magnetic decantation, and washed with water and ethanol to remove the residual substance, then redispersed in bu ffer solution for further use.

#### *2.5. Fabrication of FeOx*/*ZnS@MIPs*

The FeOx/ZnS@MIPs for BPA were prepared through molecular imprinting on the surface of magnetic fluorescence nanoparticle. For the fabrication of FeOx/ZnS@MIPs, 10 mL of a methanol solution containing 10 mg BPA and 60 μL APTES were first added to a 25 mL flask and stirred for 30 min. Then, 60 mg of the as-prepared FeOx/ZnS nanoparticle and 100 μL TEOS were added in sequence. After continuously stirring for another 30 min, 2.5 mL of 5% NH3·H2O (the catalyst) was added. The solution was deoxygenated by purging with nitrogen and stirred overnight. Non-imprinted polymers (FeOx/ZnS@NIPs) were prepared by the same procedure but without the addition of BPA. The resultant products were magnetically decanted and washed with a mixture of methanol and acetic acid (9:1, *v*/*v*) to remove the template molecules until the fluorescence intensity of FeOx/ZnS@MIPs was not changed and was similar to that of the FeOx/ZnS@NIPs ones.

#### *2.6. Fluorescence Sensing of BPA*

All of the fluorescence measurements were examined using the same condition. The excitation wavelength was set to be 311 nm with the fluorescence intensity recorded at 586 nm. The recording fluorescence emission spectrum was ranged from 350 to 700 nm.

The standard solution of bisphenol A was prepared first. Then, various samples in concentrations ranging from 1.0 to 80 ng mL−1 were made by diluting with citrate buffer solution (0.02 mol L−1, pH = 6.4). Two milligrams of FeOx/ZnS@MIPs or FeOx/ZnS@NIPs were dispersed in 10 mL testing samples. After incubating the samples for 5 min at room temperature, the changes in the fluorescence intensity of the solutions were recorded using a spectrofluorometer. This fluorescence quenching model was in accordance with the Stern–Volmer equation [48]:

$$F\_0/F = \ 1 + K\_{\rm SV} \mathcal{C}\_{\rm BPA} \tag{1}$$

where *F*0 is the initial fluorescence intensity in the absence of the BPA, *F* is the fluorescence intensity in the presence of the BPA, *K*SV is the quenching constant, and *C*BPA is the concentration of BPA.

## *2.7. Binding Selectivity*

The selectivity experiments were carried out with PTBP, BP and BPZ as structural analogs of BPA to run a batch rebinding test. Briefly, PTBP, BP and BPZ samples in the concentration ranged from 1.0 to 80 ng mL−1 and were made by diluting with citrate buffer solution. Two milligrams of FeOx/ZnS@MIPs or FeOx/ZnS@NIPs were dispersed in 10 mL testing samples and incubated for 5 min. Then the changes in the fluorescence intensity of different samples were recorded, and the imprinting factor (IF) and selectivity coefficient (SC) were used to evaluate the selectivity properties of FeOx/ZnS@MIPs and FeOx/ZnS@NIPs toward the template BPA and structural analogs [25]:

$$\text{IF } = \text{ } \lnot \text{\textquotedblleft} \lnot \text{\textquotedblright} \tag{2}$$

$$\text{SC} = \text{ IF/IF} \tag{3}$$

where *K*MIP and *K*NIP are the slopes of the linear equation of FeOx/ZnS@MIPs and FeOx/ZnS@NIPs with the target molecule, respectively, IF and IF' are the imprinting factor for template BPA and structural analogs, respectively.

#### *2.8. Analysis of Real Samples*

The prepared FeOx/ZnS@MIP was directly applied to the detection of BPA in drinking water, tap water, and lake water. The drinking water was commercial pure water purchased from the market. The tap water was collected from the laboratory. The lake water samples were collected from three different lakes (located in Beijing, China). All the samples were filtered through a 0.45 μm filter and stored at 4 ◦C.

The detection strategy was divided into two steps: (a) the enrichment and separation of BPA and (b) fluorescence detection (Figure S1). Firstly, the FeOx/ZnS@MIPs were dispersed in the water samples. The target BPA molecule was specifically bound onto the MIP layer of the FeOx/ZnS@MIPs after incubating at room temperature. Secondly, the fluorescence quench of FeOx/ZnS@MIPs in each sample was recorded and the concentration of analytes in the samples was calculated. There were

no other pretreatment procedures employed in the sample preparation. To evaluate the developed method, a recovery test was carried out by using the samples spiked with BPA standard solution.

#### **3. Results and Discussion**

#### *3.1. Synthesis of the FeOx*/*ZnS@MIPs*

The design of the FeOx/ZnS@MIPs was mainly based on coating the molecularly imprinted polymer layer on the surface of FeOx/ZnS NPs. Figure 1 illustrates the two major synthetic steps of the proposed fluorescence sensing polymer. In the first step, mercaptoethylamine was grafted onto the surface of the Mn2+-doped ZnS QDs. The mercapto of MEA was tightly bound at the surface of the bare QDs through ligand competition. The introduction of amino group to the QDs not only increases the water dispersion ability of QDs, but also provides the possibility of being combined with FeOx magnetic nanocrystals. FeOx magnetic nanocrystals were successfully synthesized by a modified solvothermal method. The –COOH was grafted onto the surface of FeOx in just one synthesis step. The carboxyl-functionalized FeOx NPs were conjugated with MEA capped QDs to prepare the multifunctional nanocomposites. The FeOx/ZnS NPs integrated the distinct properties of the optical characteristics of QDs, and the magnetic separation ability of MNPs through an EDC/NHS reaction process. In the second step, APTES was chosen as a functional monomer that had noncovalent interactions with bisphenol A [49]. The monomer (APTES) interacted with the template molecule (BPA) through a hydrogen bond to form a "pre-polymerization" complex (Figure 1). The resultant FeOx/ZnS NPs were used as substrate, TEOS and NH3·H2O were used as the crosslinker and catalyst, respectively. The pre-polymerization complex was subsequent immobilized on the surface of FeOx/ZnS NPs through a facile molecular imprinting process. The imprinting layer was coated on the FeOx/ZnS NPs to produce a "core-shell" structure. This core-shell composite provides selectivity to the template and prevents other interfering molecules from contacting the FeOx/ZnS NPs. After the removal of the template molecule BPA, the MIP layer with imprinted cavities complementary to the BPA in size, shape, and functional groups was obtained. The quantum yield (QY) of the FeOx/ZnS NPs was 20.6%, as calculated by equation S1. The resultant FeOx/ZnS@MIPs, as an ideal candidate material, was able to be used as a multifunctional sensor for high selectivity and sensitivity magnetic separation and the fluorescent detection of target BPA.

**Figure 1.** Schematic illustration of the process for the fabrication of the FeOx/ZnS@MIP-based sensor. BPA: bisphenol A.


**8. Future perspectives for Network Centric Therapy for reflex, gait, and movement disorder assessment with machine learning**

Network Centric Therapy is anticipated to have a transformative influence on the healthcare and biomedical industry. Conformal wearable and wireless inertial sensor systems are envisioned to enable historical and distinctly quantified data for subjects undergoing rehabilitation and subjects with neurodegenerative movement disorders, such as Parkinson's disease and Essential tremor. Data science methodologies can be incorporated to optimize the respective therapy strategy. With the amalgamation of machine learning and eventually deep learning conformal wearable and wireless inertial sensor systems are predicted to considerably advance augmented clinical situational awareness for diagnostic and prognostic capabilities. In particular, with the Cloud computing accessibility intrinsic to Network Centric Therapy, the most talented clinical resources from anywhere in the world can provide optimal patient specific rehabilitation and therapy to subjects from the convenience of a homebound setting. Additionally, the inherent aspects of Network Centric Therapy, such as conformal wearable and wireless inertial sensor systems, machine learning, and Cloud computing access, imply a plausible pathway to the closed-loop optimization of deep brain stimulation parameter configurations.


**Author details**

As introduced in this chapter, the use of biostimulants has potential as a powerful countermeasure for improving crop production under conditions of environmental stress. Traditional methods for enhancing yield, such as fertilizer and irrigation support, enable a basal level of production to be maintained (**Figure 4**). However, when crops are subject to unusual abiotic and/or biotic stresses, the transient use of adequate biostimulant(s) helps to overcome these stresses, sustaining production to at least the basal level, and sometimes bringing about an increase in crop production, as explained in **Figure 3**. Some biostimulants are already commercially available, and their use will become increasingly popular. However, there is still a lack of technical information for each biostimulant, such as the application period, concentration, and target plant species; these points must be established if biostimulant application is to become a reliable technique. Moreover, the combined use of biostimulants and traditional pesticides must be examined in order to realize the integrated chemical control of abiotic/biotic stress tolerance. In addition to the chemical biostimulants that are the focus of this chapter, other types of biostimulants that are derived from natural materials, such as microorganisms and algae, are also useful in reducing damage caused by abiotic stress [14, 15]. While the mode of action in chemical biostimulants can be explained scientifically, this is not the case for natural biostimulants. Therefore, more details on scientific analyses of the mechanisms used by biostimulants are necessary to support their avail-

traditional countermeasures and biostimulants are indicated by white and black arrows, respectively.

**Figure 4.** Integrated chemical control of abiotic stress tolerance using biostimulants. Traditional countermeasures, such as irrigation and fertilizers, contribute to improving the basal level of crop production. On the other hand, unusual losses in crop yield caused by biotic and abiotic stress can be mitigated by pesticides and biostimulants, respectively. In this figure, unusual adverse events and biostimulants are boxed and underlined, respectively. The stresses targeted by

A part of unpublished data introduced in this chapter was generously provided by Phyto-

ability for widespread use in the field.

140 Plant, Abiotic Stress and Responses to Climate Change

**Acknowledgements**

chrome Inc. (Tokyo, Japan).

Yasuo Yamauchi Address all correspondence to<EMAIL_ADDRESS>Graduate School of Agricultural Science, Kobe University, Kobe, Japan


**1. Introduction**

100 Ion Exchange Technologies

109.

10795.

[14] Pillay K. Nanomaterials for the Removal and Recovery of Heavy Metal Ions from

[15] Chen, C., Wang, X. Adsorption of Ni(II)f from aqueous solution using oxidized multi-

[16] Kandah, M.I., Meunier, J.L. Removal of nickel ions from water by carbon nanotubes.

[17] Lu, C., Liu, C., Rao, G.P. Comparison of sorbent cost for the removal Ni2+from aqueous solutions by carbon nanotubes and granular activated carbon. Journal of Hazardous

[18] Li, Y.H., Wang, S., Luan, Z., Ding, J., Xu, C., Wu, D. Adsorption of Cd(II) from aqueous

[19] Wang, H.J., Zhou, A.L., Peng, F., Yu, H. Chen, L.F. Adsorption characteristic of acidified carbon nanotubes for heavy metal Pb(II) in aqueous solution. Materials Science and

[20] Yang, S., Li, J., Shao, D., Hu, J. Wang, X. Adsorption of Ni(II) on oxidized multi-walled carbon nanotubes: Effect of contact time, pH , foreign ions and PAA Journal of

[21] Sheng, G., Li. J., Shao, D., Hu, J., Chen, C., Chen, Y. Wang, X. Adsorption of copper(II) on multi-walled carbon nanotubes in the absence and presence of humic or fulvic acids.

[22] Pyrzynska, K., Bystrzejewski, M. Comparative study of heavy metal ions sorption onto activated carbon, carbon nanotubes and carbon-encapsulated magnetic nanoparticles. Colloids and Surfaces A: Physiochemical and Engineering Aspects 2010; 362 pp. 102-

[23] Cech, J., Curran, S.A., Zhang, D., Dewald, J.L., Avandhandula, A., Kandadai, M., Roth S. Functionalisation of Multi-walled carbon nanotubes: Direct proof of sidewall

[24] Pillay, K., Cukrowska E.M., Coville, N.J. Improved uptake of mercury by sulphur-

[25] Nxumalo, E.N., Coville, N.J. Nitrogen Doped Carbon Nanotubes from Organometallic

[26] Perez-Aguilar, M.V., Munoz-Sandoval, E., Diaz-Florez, P.E., Rangel-Mendez, J.R. Adsorption of Cadmium and Lead onto oxidized nitrogen-doped mutiwalled carbon nanotubes: Equilibrium, and Kinetics. Journal of Nanoparticle Research, 2010; 12 467-480. [27] Aguiar, A.L., Fagan, S.B., da Silva, L.B., Mendez Filho, J., Souza Filho, A.G. Benzonitrile adsorption on Fe-doped carbon nanostructures. J. Phys. Chem. C. 2010; 114 10790-

[28] Tan, X., Fang, M., Chen. C., Yu, S., Wang, X. Counterion effects of nickel and sodium dodecylbenzene sulfonate adsorption to multiwalled carbon nanotubes in aqueous

[29] Li, Y.H., Ding, J., Luan, Z., Di, Z., Zhu, Y. Xu, C., Wu, D., Wei, B. Competitive adsorption of Pb2+, Cu2+and Cd2+ from aqueous solution by multi-walled carbon

containing carbon nanotubes. Submitted to Microchemical Journal.

solution by surface oxidized carbon nanotubes. Carbon 2003; 41, 1057-1062.

Industrial Effluents. PhD. thesis. University of the Witwatersrand; 2011.

walled carbon nanotubes. Ind. Eng. Chem. Res*.* 2006; 45 9144-9149.

Journal of Hazardous Materials 2007;146 283-288.

Materials 2008; 151 239-246.

Engineering A 2007; 466 201-206.

Hazardous Materials 2009; 166 109-116.

Journal of Hazardous Materials 2010; 178 333-340.

thiolation. Phys. Stat. Sol*.* 2006; 243 (13) 3221-3225.

Compounds: A Review. Materials 2010; 3 2141-2171.

solution. Carbon 2006; 46 1741-1750.

nanotubes. Carbon 2003; 41 2782-2792.

Rare earth elements of high purity play a significant role in many areas of contemporary techniques. They also have many scientific applications. For example, their compounds are used as catalysts in the production of petroleum and synthetic products, lanthanides are used in lamps, lasers, magnets, phosphors, motion picture projectors, and X-ray intensifying screens. The addition of the pyrophoric mixed rare-earth alloy called Mischmetal or lanthanide silicates improves the strength and workability of low alloy steels. Therefore, the preparation of high purity rare earth elements is very important for such technologies.

Mainly cation exchangers and elution by complexing agents are used for separation of rare earth elements using ion exchange methods. In this process the order of elution of individual rare earth(III) elements depends on the values of stability constants of formed complexes. They generally increase from light lanthanides(III) to heavy lanthanides(III). Ion exchange of rare earth elements in the presence of chelating ligands on anion exchangers is still a poorly studied field. However, the papers published during the last few years show particular applicability of anion exchangers to this end. The anion exchangers have been used so far for separation of thorium(IV) and uranium(IV,VI) from lanthanides(III) with mineral acid solutions and for the studies of chromatographic separation of rare earth(III) elements. As for isotopes of these elements separation processes were mainly of analytical or physicochemical character. For separation of rare earth elements the impregnating resins are also used.

In separation of rare earth(III) element complexes with chelating organic ligands strongly basic, gel polystyrene anion exchangers of quaternary ammonium groups proved to be the most effective. The research carried out dealt mainly with application of such complexing

© 2012 Kołodyńska and Hubicki, licensee InTech. This is an open access chapter distributed under the terms of the Creative Commons Attribution License (http://creativecommons.org/licenses/by/3.0), which permits unrestricted use, distribution, and reproduction in any medium, provided the original work is properly cited. © 2012 Kołodyńska and Hubicki, licensee InTech. This is a paper distributed under the terms of the Creative Commons Attribution License (http://creativecommons.org/licenses/by/3.0), which permits unrestricted use, distribution, and reproduction in any medium, provided the original work is properly cited.

agents as: ethylenediaminetetraacetic acid (EDTA), iminodiacetic acid (IDA), N' -(2 hydroxoethyl)ethylenediamine-N,N,N' -triacetic acid (HEDTA) and *trans*-1,2 cyklohexanediaminetetraacetic acid (CDTA) in lanthanides separation.

Particular attention is also paid to separation and removal of rare earth(III) elements nitrate complexes by means of frontal analysis from the polar organic solvent-H2O-HNO3 on anion exchangers of various types. The affinity series of rare earth(III) elements nitrate complexes depends on the kind of functional groups, kind of the skeleton, porosity of skeleton, cross linking degree of anion exchanger skeleton as well as kind and concentration of polar organic solvent, concentration of nitric acid, addition of another organic solvent and concentration of rare earth(III) elements.

In the paper the research on the applicability of different types of anion exchangers for the separation of rare earth elements in the presence of the complexing agents IDA, HEDTA and CDTA will be presented. The effect of the addition of a polar organic solvent (methanol, ethanol, acetone, 1-propanol, 2-propanol) on separation of rare earth(III) elements in such system will be also discussed. The examples of the removal of rare earth(III) elements nitrate complexes from the polar organic solvent-H2O-HNO3 will also be presented in detail.


*2.2. Lynx-Killed Roe Deer*

The roe deer kills of very high frequency (VHF) radio-collared and snow-tracked lynx were routinely located in the period 1995–2008 [25,26]. Field procedures are described in Nilsen et al. 2009 [25]. Permissions were granted by the Norwegian Environment Agency and procedures were approved by the Norwegian Committee for Experimental Animal Welfare (permit numbers 08/127430, 07/81885, 07/7883, 2004/48647, 201/01/641.5/FHB, 127/03/641.5/fhb, 1460/99/641.5/FBe, 1081/97/641.5/FBe, and Norwegian Institute for Nature Research (NINA) 1/95). Most of the lynx-killed roe deer carcasses included in this study were found in winter (72 in winter and 7 in summer, Table S1). Kills were checked at various times. When studying predation behavior of collared lynx we only approached kills when the lynx had abandoned the kill. When snow-tracking unmarked lynx or trying to recapture already collared lynx, we would approach a new found kill straight away in order to place traps. When the kills were examined, any signs of the presence of scavengers were recorded (such as scats, footprints, feathers) as well as signs of the kill being covered by the lynx. However, we cannot exclude that some scavengers remained undetected. We knew lynx identity for 53 lynx-killed roe deer carcasses. For this subset of data, we explored individual variation in lynx caching behavior by comparing the proportion of covered carcasses for single individuals vs. females with kittens, and for males vs. females, by means of Chi-square tests. We had information on habitat type for 66 of the lynx-killed roe deer carcasses. For this subset of data, we compared the proportion of covered carcasses in open habitats and forest habitats, by means of Chi-square tests.

We used multivariate logistic regression models to explore which variables better explained the probability of a carcass being scavenged or not. The full model explaining the probability of a carcass being scavenged included number of days after being killed, the treatment (not covered/covered) and their interaction as explanatory variables. The final model was selected by model reduction, where only explanatory variables with significance *p* < 0.05 were retained in the model.