fig1 was already discussed in connection with the description of the prior art . fig2 illustrates a solution according to the invention in the form of block diagram . it includes as functional blocks a bandpass filter 100 , phase difference detector 200 and an amplifier 250 . a signal v i with a frequency f and phase φ i is brought to the filter . signal v i may be frequency modulated , in which case f refers to its center frequency . the filter outputs a signal v o the frequency of which equals that of the input signal and the phase of which is φ o . summed to the input signal v i is noise n i and to the output signal v o noise n o . because of the filter , the power density of the output signal noise n o at a given distance from frequency f is smaller than that of the input signal noise n i . the input and output signals of the filter 100 are taken to the phase difference detector 200 which produces a signal v e proportional to the phase difference φ o − φ i . this signal is amplified in block 250 . in the real circuit , the amplifier may be included in the phase difference detector . signal v c produced by the amplifier 250 is brought to the filter 100 where it determines the location of the pass band of the filter according to the basic idea of the invention described above : feedback forces the center frequency of the pass band to follow the signal frequency f . fig3 shows an example of the frequency response of a resonator - type bandpass filter . the frequency response comprises an amplitude response 32 and phase response 31 . amplitude a is at its greatest at resonance frequency f r . phase φ , or the phase difference between the output and input signals , changes from − 90 ° to + 90 ° as the frequency increases from a very low to a very high value . at the resonance frequency f r the phase φ is zero . the invention makes use of the shape of the phase response . around the resonance frequency f r the phase φ changes relatively steeply as the frequency f changes , in other words the slope δφ / δf of the phase response 31 is rather steep , i . e . the sensitivity coefficient s 1 is relatively great . consequently , by following the phase change the resonance frequency f r can be set to the input signal frequency more accurately than by following e . g . the amplitude change . fig4 shows a linear model of a system according to the invention , corresponding to fig2 and 3 . the model includes three sensitivity coefficients s 1 , s 2 and s 3 . first , a difference δf of the input signal frequency f and resonance frequency f r is produced . said difference is multiplied in block 41 by sensitivity coefficient s 1 of the resonance circuit , producing a phase difference δφ corresponding to the frequency difference . in block 42 the phase difference δφ is multiplied by sensitivity coefficient s 2 coefficient s 2 equals the ratio δv c / δφ . so , it is indicative of how much the signal v c produced by the phase difference detector 200 and the associated amplifier 250 changes as the phase difference changes . in block 43 the change δv c of signal v c is multiplied by sensitivity coefficient s 3 . coefficient s 3 equals the ratio δf r / δv c . so , it is indicative of how much the resonance frequency f r of the resonator 100 changes as the control signal v c changes . change δf r is added to a constant frequency f ro corresponding to the center frequency of the operating frequency range , producing an instantaneous value of the resonance frequency f r . the product of the sensitivity coefficients s 1 , s 2 and s 3 equals the total gain g of the loop formed by the structure , said g being a dimensionless number . fig4 gives g ( f − f r )+ f ro = f r , so f r =( gf + f ro )/( g + 1 )≈ f −( f − f ro )/ g ( 1 ) the latter expression for the resonance frequency f r holds when the loop gain g is great . it can be seen that if the gain g is , say , one thousand , the resonance frequency f r is set at the input signal frequency f with an accuracy where the error is one per mille of the difference between frequency f and the center frequency f ro of the range . fig5 shows an example of the circuit implementation of the invention , corresponding to fig2 . it includes a bandpass filter 100 , analog multiplier 210 , amplifier stage 251 , low - pass filter 220 and a buffer amplifier 300 . the input signal of the structure is v i the frequency of which is f and phase φ i . the output signal of the whole structure is v out . the bandpass filter 100 comprises a parallel resonance circuit the inductive branch of which is formed by coil l and the capacitive branch of which is formed by capacitance diodes cd 1 , cd 2 connected in series . in addition , the bandpass filter 100 comprises a capacitor c 1 via which the input signal v i is fed to the resonator . the output signal of the filter 100 is v o the phase of which is φ o . the input signal v i is conducted via capacitor c 2 and output signal v o via capacitor c 3 to the analog multiplier 210 which gives their product v m the analog multiplier is based in this example on a widely known gilbert cell 211 which has three differential pairs q 1 - q 2 , q 3 - q 4 and q 5 - q 6 . one of the mutually multiplied signals . controls pair q 1 - q 2 and the other controls pairs q 3 - q 4 and q 5 - q 6 . the output signal v m of the analog multiplier 210 is proportional to where t represents time . phase φ e refers to the deviation from 90 ° of the phase difference of the input voltages of the gilbert cell . it equals the phase difference φ o − φ i . signal v m is brought to the amplifier stage 251 which comprises a transistor q 7 , resistor r 1 connected from the emitter of said transistor to the supply voltage v s , and resistor r 2 connected from the collector to the ground . the voltage gain is in this case r 2 / r 1 . from the collector of transistor q 7 the signal is conducted to the low - pass filter 220 . the filter comprises series - connected resistor r 3 and capacitor c 4 as well as resistor r 4 one end of which is connected to the ground . the signal is taken out between resistor r 3 and capacitor c 4 . when r 4 & lt ;& lt ; r 3 + r 2 and time constant r 4 · c 4 & gt ;& gt ; 1 /( 2 πf ), the filter attenuation at frequency 2f is ( r 3 + r 2 )/ r 4 , i . e . very high . at low frequencies no attenuation occurs , so the above - mentioned expression for signal v m is reduced down to the relatively slow - changing first term sin φ e . when the phase difference φ i − φ i is relatively small the output signal v c of the filter is proportional to it . signal v c is conducted via resistor r 5 between the resonator &# 39 ; s capacitance diodes cd 1 , cd 2 . signal v c determines the total capacitance of the diodes and thus the resonance frequency of the resonator . a suitable dc voltage level for the control voltage v c is obtained by suitable dimensioning of the amplifier stage 251 . if , for example , the frequency f of signal v i increases , the resonator produces for the phase difference φ o − φ i a positive non - zero value . the internal coupling of the analog multiplier 210 is such that the portion proportional to the phase difference in voltage v m then decreases . this results in the increase of the current of transistor q 7 and further in the increase of the control voltage v c . the increase of the control voltage v c causes the capacitance of the resonator to drop and , hence , the resonance frequency to increase . the change comes to an end when the resonance frequency has reached the input signal frequency . the output signal v o of the filter 100 is conducted via a decoupling capacitor c 5 to the buffer amplifier 300 , to the base of transistor q 8 . transistor q 8 is connected so as to function as an emitter - follower : the collector is connected directly and the base via resistor r 6 to the supply voltage line , and the emitter via resistor r 7 to the ground . the output signal v out of the whole structure is taken from the emitter of transistor q 8 via a decoupling capacitor c 6 . it also follows from the feedback described above that the phase φ o follows the phase φ i of the input signal when changes occur in the latter while the frequency f stays constant . in this case the resonance frequency changes momentarily until the phase difference is reduced to zero . independent of the phase difference follow - up , the filter 100 can be on the other hand easily dimensioned such that signal v o follows relatively low - frequency amplitude changes that possibly occur in the input signal v i . fig6 shows another example of the circuit implementation of the invention . the filter 100 is a bulk acoustic wave ( baw ) resonator . the filter &# 39 ; s input and output signals are marked as in fig5 . they are in this example mixed using a well - known balanced diode mixer 215 which comprises matched diodes d 1 , d 2 , d 3 and d 4 as well as transformers t 1 and t 2 . when the frequency components the frequencies of which are equal to or greater than the input signal frequency f are removed from the output signal v m of the mixer 215 , a signal is left which is proportional to the phase difference φ o − φ i . said high - frequency components are attenuated by an amplifier unit 240 . it includes an operational amplifier a 1 connected as an inverting amplifier . the input impedance of the operational amplifier is r 1 and the feedback impedance comprises a parallel connection one branch of which comprises resistor r 2 and the other branch of which comprises resistor r 3 and capacitor c 3 connected in series . the gain of unit 240 at relatively low frequencies is then r 2 / r 1 , and at relatively high frequencies ( r 2 ∥ r 3 )/ r 1 , where r 2 ∥ r 3 represents the resistance of the parallel connection of resistors r 2 and r 3 . in general the letter “ r ” in the drawing refers to both a resistor and its resistance . similarly , “ c ” refers to both a capacitor and its capacitance . the desired filtering effect is achieved when r 3 & lt ;& lt ; r 2 . the output signal v c of unit 240 is brought to the baw resonator 100 where it determines the resonance frequency of the resonator . the output signal v o of the filter 100 is brought to a similar buffer amplifier 300 as in fig5 in order to produce the output signal v out . above it was described solutions according to the invention , but the invention is not limited to the solutions described . for example , it is possible to use other known structures for the filter and the circuit that follows the phase difference . the invention can also be used in connection with systems that use different modulation and coding schemes , such as e . g . tdma ( time division multiple access ), cdma ( code division multiple access ) and umts ( universal mobile telecommunications system ). the inventional idea can be applied in many different ways within the scope defined by the independent claims .