referring to fig1 photodetector circuit 1 detects light 6 impinging on photodiode 5 , producing a photocurrent therein . such photocurrent flows through the feedback network 9 to thereby produce a corresponding output voltage v out on conductor 26 . conductor 26 is the output of a unity gain buffer amplifier 8 , the input of which is connected to conductor 33 . conductor 33 is connected to the output of transimpedance amplifier 2 . the basic structure shown in fig1 including a photodiode , a transimpedance amplifier , and a buffer circuit , is well known in the prior art . transimpedance amplifier 2 includes a current mirror circuit including a current source 21 connected to the base and a collector of a pnp control transistor 20 having its base and collector also connected to the bases of pnp constant current source transistors 14 , 18 and 22 , which have their emitters coupled to v +. transistor 14 has its collector connected to the emitters of emitter - coupled pnp differential amplifier input transistors 10 and 11 . the collectors of pnp transistors 10 and 11 are connected to the collectors of npn load transistors 12 and 13 , respectively , which form an npn current mirror . the collector of transistor 12 is connected to the bases of transistors 12 and 13 , the emitters of which are connected to ground . the collector of transistor 13 is connected to the base on npn transistor 15 , the emitter of which is connected to ground , and the collector of which is connected to conductor 33 . conductor 33 is connected to the collector of pnp current mirror output transistor 22 . buffer circuit 8 includes npn pull - up transistor 23 having its collector connected to v +, its emitter connected to output conductor 26 , and its base connected to conductor 33 . pnp pull - down transistor 24 has its base connected to conductor 33 , its emitter connected to conductor 26 , and its collector connected to ground . however , in the photodetector circuit 1 of fig1 the input bias current sources 40 and 41 of the prior art circuit in fig2 are omitted , and bypass capacitors 16 and 17 are connected between the base and emitter of pnp input transistor 31 and pnp input transistor 32 , respectively . the elimination of bias current sources 40 and 41 of the prior art circuit fig2 significantly reduces the input bias current of the circuit because instead of having the previously mentioned bias currents of 1 to 50 microamperes flowing through pnp input transistors 31 and 32 as in the prior art circuit of fig2 only the minute base currents of input pnp transistors 10 and 11 flow through the emitters of pnp input transistors 31 and 32 , respectively . such base currents are only approximately 50 nanoamperes , and result in very high dynamic emitter resistance r e are of transistors 31 and 32 . nevertheless , the circuit of fig1 achieves approximately the same good &# 34 ; low noise performance &# 34 ; as the prior art circuit fig2 . since the quiescent current in input transistors 31 and 32 is substantially reduced compared to the bias or quiescent current in the corresponding input transistors in prior art fig2 the frequency response of pnp input transistors 31 and 32 is degraded . however , bypass capacitors 16 and 17 feed the input signals from input conductors 3 and 4 directly to emitter - coupled pnp input transistors 10 and 11 , thereby providing good frequency response for the circuit . furthermore , bypass capacitors 16 and 17 improve input referred noise characteristics of the circuit . the result of the circuitry shown in fig1 with bias current sources 40 and 41 in the prior art circuit of fig2 omitted and bypass capacitors 16 and 17 provided as shown , provides a transimpedance amplifier that can operate from a single power supply , has low quiescent bias currents in the input transistors , operates with low noise , and has an input common mode voltage range that includes ground . such good low noise performance is achieved by the circuit of fig1 even though the base currents of pnp input transistors 10 and 11 include large noise current components . without bypass capacitors 16 and 17 , such large noise current components would flow through the emitters of pnp input transistors 31 and 32 , respectively , and hence would develop large noise voltages across the large dynamic emitter resistances r e of pnp input transistors 31 and 32 . ( the dynamic emitter resistances r e are large because they are inversely proportional to the small base currents of transistors 10 and 11 which serve as bias currents in the emitters of pnp input transistors 31 and 32 . in contrast , in the prior art circuit of fig2 the bias currents applied by current sources 40 and 41 are very large and therefore produce low values of r e in transistors 31 and 32 , resulting in low differential noise voltages between the bases of pnp differential amplifier input transistors 10 and 11 and hence its above mentioned low - noise operation .) the high values of r e of pnp transistors 31 and 32 ordinarily would tend to produce very large differential noise voltages between the bases thereof . however , at high frequencies bypass capacitors 16 and 17 provide very low impedances connected in parallel with the high dynamic emitter resistances r e of pnp input transistors 31 and 32 . therefore , at the high frequencies of most interest ( e . g ., 1 kilohertz and beyond ), the noise current components of the base currents of pnp transistors 10 and 11 flow through the low impedances of bypass capacitors 16 and 17 , respectively , and consequently the noise voltages generated between the bases of pnp transistors 10 and 11 are very low . therefore , the high differential noise voltages which ordinarily would result from the high values of r e of transistors 31 and 32 are greatly reduced in the circuit of fig1 while the low input bias current is very low as a result of the very low bias currents of transistors 31 and 32 . the preferred embodiment of transimpedance amplifier 2 is shown in the circuit of fig6 and sums a &# 34 ; ptat current &# 34 ; ( proportional to absolute temperature ) with a &# 34 ; v be / r current &# 34 ; to provide a current in constant current source transistor 18 with a temperature coefficient ( t / c ) near zero . the circuitry in fig1 includes constant current source transistor 18 connected to supply a current i 1 flowing through a resistor 19 to ground . the junction between the collector of constant current source transistor 18 and resistor 19 is connected to the (+) input conductor 4 of transimpedance amplifier 2 . the value of the current i 1 and the value of resistor 19 are selected so that the bias voltage across resistor 19 , referred to herein as a &# 34 ; pedestal voltage &# 34 ;, is equal to the maximum expected input offset voltage of transimpedance amplifier 2 , i . e ., approximately 10 millivolts . typical variations in the v be voltages of pnp input transistors 10 and 11 and pnp input transistors 31 and 32 result in the sum of the input offset voltage ( v offset ) of transimpedance amplifier 2 and the voltage developed across the feedback resistor r + by the input bias current being typically in the range from 5 to 10 millivolts . the reason the &# 34 ; pedestal voltage &# 34 ; v bias is desirable can be explained with reference to fig1 a . with no incident light impinging on photodiode 5 , the only current flowing through photodiode 5 is the so called &# 34 ; dark current &# 34 ; i dark . an input bias current i bias flows out of the (-) input of transimpedance amplifier 2a . if transimpedance amplifier 2a has a pnp differential input stage , the common mode voltage range includes ground , so a feedback current i f will flow through feedback resistor r f in the direction indicated in fig1 a to equalize the voltages on the inverting and non - inverting inputs of transimpedance amplifier 2a . typically , i bias will be a few nanoamperes , r f typically is a few megohms , and the input offset voltage of v os of transimpedance amplifier 2a will be a few millivolts . this will tend to cause the output voltage v out on conductor 2c to swing below ground . however , the output of a typical single - supply transimpedance amplifier 2a , is not capable of swinging below ground . consequently , a nonlinearity in amplification exists when v out is near ground . the addition of a few millivolts of a &# 34 ; pedestal &# 34 ; bias voltage as described above compensates for the input offset voltage of transimpedance amplifier 2a and the voltage developed across the feedback resistor by the input bias current , and therefore ensures linear operation when v out is near ground . without constant current source transistor 18 and resistor 19 , i . e ., with conductor 4 connected to ground , transimpedance amplifier 2 is likely to be completely unresponsive to the input light signal 6 impinging on photodiode 5 until the light intensity is such that the magnitude of the resulting photocurrent becomes large enough to overcome the 10 millivolt offset voltage . stated differently , there is a large &# 34 ; dead band &# 34 ; of approximately 10 millivolts in the closest prior art transimpedance amplifiers in photodetector circuits . as mentioned above , the closest prior art single power supply photodetector circuits which are designed to operate from a single power supply do not attempt to solve this problem . the input bias circuit including constant current source transistor 18 and resistor 19 selected to maintain the (+) input of transimpedance amplifier 2 at 10 millivolts above ground avoids the above mentioned &# 34 ; dead band &# 34 ; and provides linear detection and amplification of low intensity light signals . the linearity of the circuit operation can be further improved by provision of jfet 27 with its drain connected to ground as shown in fig1 so it draws a minimum amount of current through the emitter of output pull - up transistor 23 in buffer circuit 8 . this , in combination with the 10 millivolt &# 34 ; pedestal &# 34 ; bias offset voltage superimposed on the (+) input conductor 4 helps provide the desired linear amplification and allows output current sinking of typical bias currents of additional signal processing circuits that may be connected to the output of photodetector circuit 1 . apart from the advantages of jfet 27 in combination with the 10 millivolt &# 34 ; pedestal &# 34 ; bias offset voltage described above , jfet 27 can be very useful if the transimpedance amplifier 9 and buffer circuit 8 are operated between v + and ground , and it is desirable that v out be pulled all the way to ground . pnp pull - down transistor 24 can only pull v out within a v cesat of ground . for example , a simple charge pump circuit could be included on the same circuit which would apply a voltage to conductor 28 of jfet 27 so as to pull v out all the way to ground . in fig1 p - channel jfet 27 has its source and gate electrodes connected to v out conductor 26 and its drain 28 connected to an external pull - down voltage v pd that can be below ground . this feature is important in some applications wherein it is desirable to be able to connect a load supplied by v out to a power supply or pull - down voltage which is more negative than ground , especially in transimpedance amplifiers designed to operate from a single power supply , so the device can be conveniently operated with dual power supplies . pull - down jfet 27 acts as a constant current source when v pd is more negative than v out minus the threshold voltage of jfet 27 , thereby providing the desired negative pull - down current without overloading output transistor 23 , or dissipating excessive power . fig3 and 3a show the construction of photodiode 5 which is integrated into the same chip as transimpedance amplifier 2 and output buffer 8 . photodiode 5 includes a shallow p + ion - implanted region 46 which forms a photo junction 46a with the underlying n - epitaxial region 51 which is junction - isolated from other n - epitaxial regions by an annular p + isolation ring 57 . light 6 impinging on this structure produces a first photocurrent in photo junction 46a . p + contact diffusions 47 provided convenient coupling ( e . g ., by a bias voltage source ) of the p + anode of photodiode 5 to the ground or the (+) input of transimpedance amplifier 2 . an n + buried layer region 44 is provided directly under p + implanted region 46 . an n + region 48 provides for convenient connection of the cathode of the photodiode to (-) input of transimpedance amplifier 2 . in accordance with the present invention , an annular &# 34 ; guard ring tub &# 34 ; 51a of the n - epitaxial layer partly or completely surrounds the photodiode structure including p + implanted region 46 and the junction - isolated epitaxial region 51 in which region 46 is formed . n + region 49 is utilized to electrically connect n - guard ring tub 51a to the positive power supply v +, lightly doped p - type substrate 50 being connected to ground via a p + contact region 57a formed in isolation ring 57 . the components of transimpedance amplifier 2 and buffer amplifier circuit 8 are included in additional junction - isolated n - epitaxial regions such as 62 which are isolated from n - guard tub 51a by a p + isolation ring 60 . in operation , the impinging light 6 to be detected strikes the entire upper surface of the integrated circuit in which photodiode 5 , transimpedance amplifier 2 , and output buffer 8 are fabricated . such impinging light 6 may include infrared or near infrared components , indicated by 6b in fig3 . infrared or near infrared light , unlike ultraviolet light , penetrates deeply into the relatively thick p - type substrate 50 , producing electron - hole pairs such as 43 . since p - type substrate 50 is lightly doped , such electron - hole pairs 43 have long lifetimes , and therefore form electron currents which are &# 34 ; collected &# 34 ; by positive biased n - guard tub 51a , instead of being collected by various n - epitaxial regions in which the bases of amplifier input transistors 10 , 11 , 31 and 32 or the like are formed , or instead of being collected by the n - epitaxial region 51 of photodiode 5 to add on an error to the photocurrent generated thereby . therefore , the structure shown in fig3 results in much more accurate detection and amplification of the detected light 6 than the prior art . an additional advantage of the structure shown in fig3 is that the n + buried layer region 44 and the lightly doped p - type substrate actually form a second photo junction 44a which is , in effect , connected in parallel with the photo junction 46a of the &# 34 ; top &# 34 ; photodiode formed by between implanted p + region 46 and n - epitaxial region 51 . the photo junction 44a of the &# 34 ; bottom &# 34 ; photodiode formed by n + buried layer region 44 and p - type substrate 50 is quite responsive to infrared or near infrared light , and therefore produces a second photocurrent that is summed with the first photocurrent generated by the photo junction 46a between p + implanted region 46 and n - epitaxial region 51 . the summed first and second currents then flow into n + contact region 48 from the feedback network 9 . fig3 b schematically shows the &# 34 ; composite &# 34 ; photodiode 5 including &# 34 ; top &# 34 ; photodiode 5a and &# 34 ; bottom &# 34 ; photodiode 5b connected in parallel between the (+) and (-) inputs of transimpedance amplifier 2 . in fig3 b , numeral 6b designates the impinging infrared or near infrared component of the detected light 6 , and numeral 6a designates impinging light of all other wavelengths . it has been found that the structure of fig3 provides a superior photodiode that is responsive to a much broader range of ambient light wavelengths than the closest prior art of which the inventors are aware , which prior art applies the photocurrents provided by the &# 34 ; top &# 34 ; and &# 34 ; bottom &# 34 ; photodiodes to the inputs of two separate transimpedance amplifiers , respectively . in another embodiment of the invention , the feedback circuit 9 shown in fig2 in which both the feedback capacitor c f and the feedback resistor r f are connected between the (-) input 3 of transimpedance amplifier 2 and the output 26 of buffer circuit 8 , is modified as shown in fig4 . in fig4 r f is connected the same as in fig2 but c f is connected between the output 33 of transimpedance amplifier 2 and the (-) input thereof . the improvement shown in fig4 and 4a - c , including connecting the drain of jfet 27 to ground , improves the previously mentioned &# 34 ; noise effective power &# 34 ; ( nep ) of an optical detector circuit . it can be shown that the effect of compensation of a feedback loop to a node ( e . g ., node 33 ) which has variable gain is to provide closed loop bandwidth and noise gain which are controlled by the gain of the output buffer 8 . in the circuit shown in fig4 the gain of output buffer 8 , which includes npn transistor 23 , pnp transistor 24 , and jfet 27 , is controlled as a function of the output voltage v out . if the gain of output stage 8 is reduced , both the &# 34 ; noise gain &# 34 ; and the open loop bandwidth are reduced . this can be intuitively understood by recognizing that the feedback capacitance c f is effectively being multiplied by the inverse of the gain of buffer circuit 8 . that results in reduced output noise , thereby reducing ( and hence improving ) the noise effective power nep . the equivalent circuit of the dynamic emitter resistance r e of transistor 23 and the &# 34 ; on &# 34 ; resistance r on of jfet 27 is shown in fig4 a . it can be readily seen that as v out increases , r e rapidly falls , and the gain of the emitter follower transistor 23 increases from less than 0 . 5 to 1 as v out increases from zero to beyond about 0 . 2 volts . fig5 shows the noise gain and open loop gain as functions of output stage gain . notice that for values of output stage gain g less than 1 , both noise gain and open loop bandwidth are reduced . this results in lower output noise voltage . note that the output noise voltage is proportional to the area bounded by the noise gain and open loop curves . the lower output noise voltage effectively lowers the nep . fig4 b illustrates the values of r on and r e as a function of v out and also illustrates the ratio of r e / r on verses v out . another intuitive way of understanding why the circuit of fig4 reduces the bandwidth at low values of v out so as to substantially improve the noise performance is to recognize that low values of gain of buffer stage 8 cause the capacitance c f to appear to be much larger in value , so it essentially acts as a filter that reduces noise from the input stage . fig4 c shows the gain of buffer circuit 8 as a function of v out . specifically , for very low values of v out , the gain of output buffer 8 is considerably less than 1 , and approaches 1 as v out exceeds 0 . 2 volts . this occurs because when v out is near zero volts , the drain current through jfet 27 is nearly zero . consequently , the emitter current of npn transistor 23 is also nearly zero , so its dynamic resistance r e is very high . while the invention has been described with reference to several particular embodiments thereof , those skilled in the art will be able to make the various modifications to the described embodiments of the invention without departing from the true spirit and scope of the invention . it is intended that all combinations of elements and steps which perform substantially the same function in substantially the same way to achieve the same result are within the scope of the invention . for example , the substrate 50 in fig3 could be n - type instead of p - type , and the conductivity types of the other layers illustrated in this structure could be reversed . furthermore , the photodiode structure shown in fig3 could be provided on an integrated circuit wafer with circuitry other than transimpedance amplifier 2 and output buffer 8 . for example , the photodiode structure could be connected to a comparator or to the input of an analog - to - digital converter , voltage - to - frequency converter circuit , or other signal processing circuitry integrated onto the same semiconductor chip as the photodiode structure 5 . the constant current source pnp transistor 14 in fig1 could be replaced by a jfet with its gate and source connected to the emitters of input transistors 10 and 11 and a drain coupled to v +. the feedback resistor r f can be connected to a different conductor than v out as long as the conductor carries a voltage which represents reduced gain of the output buffer as v out decreases to values very close to ground . in the structure shown in fig3 a , additional photodiodes such as the one shown in junction - isolated n - type epitaxial region 51 could also be included in one or more of the n - type junction - isolated epitaxial regions 62 . the disclosed guard tub structure could be useful on a chip that includes an array of such photodiode structures , with no other circuitry being integrated onto the same chip . in such a photodiode array , the positively or zero biased n - type guard tubs would reduce cross talk between photodiodes .