embodiments of the present invention will be described with reference to the drawings . first , the theory of the present invention will be described for a better understanding thereof . to control the output voltage waveform as desired , an inverter must be driven by a constant current . to achieve this an inductance ( rfc ) is inserted between at least one of the input terminals of the inverter and a d . c . power supply . in this case , the output voltage waveform is determined by the input current waveform ( a rectangular wave produced by the switching of d . c . current by the inverter ), a load , and the impedances in the circuit . fig4 shows an equivalent circuit of an inverter device , the circuit impedance of which ( including c oss ) is concentrated at a load end . in the figure , rfc denotes a choke coil , and q1 to q4 denote switching elements for which , in practice , transistors are used . as indicated by l , r , and c , impedances in the inverter circuit are equivalently concentrated at a load end ; c includes the source - drain capacitance c oss ; i denotes an input current ; and v ( t ) denotes a voltage at the output terminal . fig5 shows the voltage across the load and the waveform of the current flowing to the entire load . the voltage v ( t ) across the load is also applied across the mos transistors which are in an off state , and accordingly , the waveform of the source - drain voltage of the mos transistors is approximately identical to that of the output voltage . according to studies by the inventors , the output voltage waveform can be approximately expressed by the following formulae : ## equ1 ## where i : input d . c . current value where t is in the range of 0 ≦ t ≦( 1 / 2f ) provided t = 0 at the rise time of the rectangular current wave , and thus there is a repetition of a general impulse response waveform of a secondary factor . as is clear from the above formulae ( 1 ), ( 2 ) and ( 3 ), by suitably selecting the values of l , c and r , it is possible to change the output voltage waveform to a great extent . the voltage value after a half - period from the inversion of the current waveform due to the switching of the transistors can be expressed as v 0 ( τ ), by using the aforesaid v 0 ( t ), where τ = 1 / 2f . in this case , the energy accumulated in the c oss of the transistors which are in an off state is which is consumed in the transistors when they are next turned on . at this time , a current flows to the transistors , which have been made on to charge the c oss thereof to an equal voltage level , and thus a loss occurs . this energy is similarly given by accordingly , the overall loss of the circuit at each switching is ## equ2 ## since the loss occurs twice in each period , then the total loss pl is where n is the number of mos transistors connected in parallel . as is clear from this formula , by reducing the source - drain voltage v ( τ ) when switching the transistors to 0 , pl can be made zero . fig6 shows changes of the output voltage waveform with respect to l , c and r , wherein the above conditions are fulfilled . the input power is equal for each waveform , and thus the smaller the value ζ , the smaller the peak voltage value of the waveform , and accordingly , the input electric power can be increased . the limit is ζ = 0 where the waveform is a sine wave . this means r =∞, and electric power cannot be supplied to the load . therefore , in practice , ζ is set to a relatively small value . according to experiments , when ζ = 0 . 3 , the waveform can be regarded almost as a sine wave , and a large increase in the input electric power ensured . as seen from the above , the optimum value for l , c , and r can be obtained from ## equ4 ## unrealistically the condition is expressed using the equality sign , and thus an allowable range must be provided . generally , to ensure the efficiency of the inverter , about 80 % is permissible . in this case , a permissible range for the difference of the switching timing , i . e ., the difference between the intrinsic frequency √ 1 - ζ 2 fn of the waveform and the switching frequency f , is obtained . the output power p o obtainable at load r is expressed as follows , using the formula ( 1 ), ## equ5 ## from the formula ( 7 ), the value of the loss is at a maximum when a total c is given by c oss , then where ## equ6 ## and if the loss rate ξ is defined as a reciprocal of the efficiency , then ξ is a function of only ζ and ## equ7 ## fig7 shows the relationship between k and the loss rate . since k is an index of the above - mentioned frequency difference , k and the loss rate ξ are plotted along the axes of abscissa and ordinate , respectively . therefore , under the conditions ## equ8 ## a practical maximum output can be obtained . fig1 shows a first embodiment of the present invention . in the figure , a d . c . power supply es is applied to the input terminals . rfc denotes a choke coil , and q1 , q2 , q3 and q4 denote n - channel mos transistors , to thereby constitute an inverter circuit . gate signals v gs1 , v gs2 , v gs3 and v gs4 are respectively applied to the gates of the transistors . note , v gs1 and v gs4 are the same signal and v gs2 and v gs3 are the same signal . when the transistors q1 and q4 are simultaneously turned on a forward current flows through a load lo , and when the transistors q2 and q3 are turned on , a reverse current flows through the load lo . the frequency of the current flowing through the load lo is determined by the frequencies of the gate signals v gs1 , etc . c oss1 , c oss2 , c oss3 , and c oss4 represent the drainsource output capacitances of the transistors q1 , q2 , q3 , and q4 , respectively , and l represents the inductance selected to be in a parallel resonance with the impedance , as observed from the inverter side of the load lo , to fulfill the aforesaid formulae ( 15 ). lo denotes the load , which may be a laser tube of a high - frequency pumping co 2 laser , a high - frequency heat furnace , or the like . the net resistance value of the load is r . fig2 shows a second embodiment in which a capacitor c is added at the output end impedance . when the value of the capacitor c is small as in the first embodiment and the formulae ( 15 ) are not fulfilled , a capacitor c is connected in parallel with the inductance l , to thereby fulfill the formulae ( 15 ). fig3 shows a third embodiment not using an additional capacitor c , as in the second embodiment , but in this embodiment , capacitors c1 to c4 are connected between the source and drain of the respective transistors q1 to q4 . namely , since the influence of the stray impedance in the inverter circuit is large , a resonance is obtained between both ends of each transistor ; if not , the waveforms at both ends of the transistors will not be improved . instead of adding the capacitors c1 , etc . to the respective transistors , the number of parallel connections of the transistors may be changed to adjust the impedance so as to fulfill the resonance condition . as described above , according to the present invention , the parallel resonance condition of a current inverter is adjusted to optimize the output voltage waveform , permit an increase in the input electric power , and minimize the loss , whereby the output power of the inverter is remarkably increased .