in the following description of the preferred embodiment , reference is made to the accompanying drawings which form a part hereof , and in which is shown by way of illustration a specific embodiment in which the invention may be practiced . it is to be understood that other embodiments may be utilized and structural changes may be made without departing from the scope of the present invention . referring to fig1 a transceiver 100 in a digital subscriber loop is illustrated comprising a line driver 101 for driving the digital subscriber loop , an echo canceler 102 , a decision feedback equalizer 104 for reducing postcursor intersymbol interference , a first summing means 106 , a second summing means 108 , a third summing means 110 , a digital to analog converter 114 , a comparator 116 , a first impedance network 118 , a second impedance network 120 , low pass filter means 122 , a phase equalizer 124 for reducing precursor intersymbol interference and a transformer 126 for isolating and coupling the transceiver 100 to a twisted pair cable 103 . data transmission occurs by applying a transmit signal to node 104 of the line driver 101 . the output node 107 of line driver 101 is coupled to second impedance network 120 and to the transformer 126 through a first impedance network 118 . transformer 126 electrically isolates and couples node 105 to the twisted pair cable 103 . data reception occurs by receiving a signal that is impressed from a remote location onto the twisted pair cable 103 . the receive signal impressed on twisted pair cable 103 is isolated and coupled to node 105 through transformer 126 . node 105 comprises an electrical junction between a first end of first impedance network 118 , a first input of first summing means 106 and a first terminal from the primary side of transformer 126 . a second terminal from the primary side of transformer 126 is referenced to a common voltage reference . first impedance network 118 is selected to have an impedance value close to a line impedance reflected back by transformer 126 . assuming first impedance network 118 is substantially equal to the reflected line impedance , the voltage at node 105 is substantially one - half the voltage at line driver node 107 when no signals are impressed on twisted - pair cable 103 . second impedance network 120 divides down the signal at line driver node 107 providing a signal at node 109 which is one - half that at node 107 . node 109 is coupled to a second input of first summing means 106 . first summing means 106 combines the signals from node 105 and node 109 in a subtractive relation and outputs a difference signal that is substantially free from transmit signal information at node 111 . in the ideal case when first impedance network 118 exactly equals the reflected impedance from transformer 126 , output node 111 of first summing means 106 represents only the received signal component of the signal at node 105 . node 111 is coupled to low pass filter 122 which provides a output at node 113 substantially free of high frequency components . the output node 113 is coupled to phase equalizer 124 for reducing precursor intersymbol interference . phase equalizer output node 115 is coupled to a first input of second summing means 108 . a second input to second summing means 108 is coupled to a digital to analog ( d / a ) converter 114 output node 117 . the second summing means 108 output node 121 is coupled to comparator 116 which quantizes the analog signal at node 121 into a digital signal suitable for digital processing at node 119 . node 119 is also coupled to a decision feedback equalizer 104 for postcursor intersymbol interference reduction . the output node 123 of the decision feedback equalizer 104 is digitally combined with the output node 125 of echo canceler 102 in third summing means 110 . the digital output node 127 of third summing means 110 is coupled to d / a converter 114 for conversion to an analog signal at node 117 . input node 104 is coupled to echo canceler 102 . echo canceler 102 is not essential to the understanding of the present invention , but is described in commonly assigned u . s . pat . no . 5 , 084 , 866 , issued jan . 28 , 1992 to kenneth g . buttle entitled &# 34 ; transversal filter echo canceller &# 34 ;, and co - pending u . s . application ser . no . 07 / 507 , 595 , filed apr . 10 , 1990 , to kenneth g . buttle , et . al ., entitled &# 34 ; non - linear echo canceller &# 34 ;, both of which applications are herein incorporated by reference . decision feedback equalizer 104 is well known in the art , and an understanding of it is also not essential to the present invention . an exemplary , but not exclusive , decision feedback equalizer is described in u . s . pat . no . 4 , 789 , 994 issued dec . 6 , 1988 , to randall et al . entitled &# 34 ; adaptive equalizer using precursor error signal for convergence control &# 34 ; , which patent is herein incorporated by reference . referring now to the schematic illustrated in fig2 a phase equalizer 124 implemented as a switched capacitor filter is shown . the z - transfer function t ( z ) of phase equalizer 124 has the following form : ## equ4 ## where : ## equ5 ## capacitors c1 , c2 , c3 , and c4 in fig2 are realized with unit and fractional capacitors . referring to fig3 the value of zero 202 corresponds to the a equation and the value of pole 200 corresponds to the b equation . the desired z - transfer function is determined by opening and closing switches sw1 through sw4 at a particular frequency along with selecting the values of capacitors c1 , c2 , c3 , and c4 . the relative values with respect to the unit circle 204 of pole 200 and zero 202 were found by experimentation on actual telephone lines and software simulations of worst case line conditions . pole 200 and zero 202 were selected so as to minimize precursor intersymbol interference and maximize signal - to - noise ratio under the constraint of a zero crossing point occurring at baud period (- t ). in other words , the zero crossing point of the pulse directly coincides with a precursor sample . for a worst case loop condition , the pole 200 was found to have a numerical value 0 . 689 , the zero 202 was found to have a numerical value of 0 . 477 and a gain ( g ) having a numerical value of 0 . 839 while opening and closing switches sw1 , sw2 , sw3 , and sw4 at a frequency four times the data baud . the corresponding capacitor values are : c1 = 5 . 145 ( c 0 ), c2 = 10 . 79 ( c 0 ), c3 = 4 . 0 ( c 0 ), and c4 = 8 . 861 ( c 0 ); where c 0 is a unit capacitor . fig4 illustrates an input and a corresponding phase equalizer 124 output . precursor dip 300 is induced by phase equalizer 124 to facilitate zero crossing detection at baud period (- t ). fig5 illustrates amplitude and phase characteristics in the continuous frequency domain of a phase equalizer 124 . the detailed description of the preferred embodiment of the invention having been set forth herein for the purpose of explaining the principles thereof , it is known that there may be modifications , variation or change in the invention without departing from the proper scope of the invention and the claims thereto .