the density and speed of modern vlsi technology can be applied to overcome the i / o bottleneck they have created by building sophisticated i / o circuitry that compensates for the characteristics of the physical interconnect and cancels dominant sources of timing and voltage noise . such optimized i / o circuitry is capable of achieving i / o rates an order of magnitude higher than those commonly used today while operating at lower power levels . a system embodying the invention can achieve a 4 gbps signaling rate by controlling and compensating for characteristics of the transmission medium , by cancelling timing skew , and through careful management of time and voltage noise . fig1 shows one channel of high - speed signaling system embodying the invention . a transmitter module 22 accepts 8 - bit parallel data at 400 mhz . each byte is coded into ten bits for band - limiting and forward error correction and transmitted across a single differential communications channel such as a transmission line , a cable , a circuit board trace or an optical fiber . the lossy transmission line as well as package and connector parasitics attenuate and distort the received waveform , and it is further corrupted by noise coupled from adjacent lines and the power supply . the receiver 24 accepts this noisy , distorted signal and its own 400 mhz clock . the receiver generates 4 ghz timing signals aligned to the received data , samples the noisy signal and equalizes it in the analog domain , decodes the signal , and produces synchronous 8 - bit data out . the availability of 4 gbps electrical signaling will enable the design of low - cost , high - bandwidth digital systems . the wide , slow buses around which many contemporary digital systems are organized can be replaced by point - to - point networks using a single , or at most a few , high - speed serial channels resulting in significant reduction in chip and module pinouts and in power dissipation . a network based on 400 mbytes / s serial channels , for example , has several times the bandwidth of a 133 mbytes / s pci - bus that requires about 80 lines . also , depending on its topology , the network permits several simultaneous transfers to take place at full rate . a group of eight parallel channels would provide sufficient bandwidth ( 3 . 2 gbytes / s ) for the cpu to memory connection of today &# 39 ; s fastest processors . for modest distances ( up to 30 m with 18 awg wire ), high - speed electrical signaling is an attractive alternative to optical communication in terms of cost , power , and board area for peripheral connection and building - sized local - area networks . skin - effect resistance and dielectric loss causes the attenuation of a conventional transmission line to increase with frequency . with a broadband signal , as typically used in digital systems , the superposition of unattenuated low - frequency signal components with attenuated high - frequency signal components causes intersymbol interference that degrades noise margins and reduces the maximum frequency at which the system can operate . this effect is most pronounced in the case of a single 1 ( 0 ) in a field of 0s ( 1s ) as illustrated in fig2 a and b . the figures show a 4 gb / s signal ( fig2 a ) and the simulated result of passing this signal across 3 m of 24 awg twisted pair ( fig2 b ). the highest frequency of interest ( 2 ghz ) is attenuated by − 7 . 6 db ( 42 %). the unattenuated low - frequency component of the signal causes the isolated high - frequency pulse to barely reach the midpoint of the signal swing , providing almost no eye opening in a differential system and very little probability of correct detection . the problem here is not the magnitude of the attenuation , but rather the interference caused by the frequency - dependent nature of the attenuation . the high - frequency pulse has sufficient amplitude at the receiver for proper detection . it is the offset of the pulse from the receiver threshold by low - frequency interference that causes the problem . the use of a receiver equalizer to emphasize the high - frequency components of the signal eliminates this problem . first , we characterize the nature of this attenuation in more detail . fig3 a – d show the resistance per meter and the attenuation per meter as a function of frequency for a 30 awg ( d = 128 mm ) twisted pair with a differential impedance of 100 ohms ( fig3 a and 3b ) and for a 5 mil ( d = 125 mm ) half - ounce ( 0 . 7 mil thick ) 50 ohms ( fig3 c and 3d ) stripguide . for the 30 awg pair , the skin effect begins increasing resistance at 267 khz and results in an attenuation to 56 % of the original magnitude (− 5 db ) per meter of cable at our operating frequency of 2 ghz corresponding to a bit rate of 4 gb / s . skin effect does not begin to effect the 5 mil pc trace until 43 mhz because of its thin vertical dimension . the high dc resistance ( 6 . 8 ohms / m ) of this line gives it a dc attenuation of 88 % (− 1 . 2 db ). above 70 mhz , the attenuation rolls off rapidly , reaching 40 % (− 8 db ) at 2 ghz . the important parameter , however , is the difference between the dc and high - frequency attenuation which is 45 % (− 6 . 8 db ). the effect of frequency dependent attenuation is graphically illustrated in the eye - diagrams of fig4 a – c . as shown in the waveform in fig4 a , without equalization , a high - frequency attenuation factor of a reduces the height of the eye opening to 2a - 1 with the eye completely disappearing at a ≦ 0 . 5 . this height is the amount of effective signal swing available to tolerate other noise sources such as receiver offset , receiver sensitivity , crosstalk , reflections of previous bits , and coupled supply noise . because the waveforms cross the receiver threshold offset from the center of the signal swing , the width of the eye is also reduced . as illustrated in fig4 b , the leading edge of the attenuated pulse crosses the threshold at the normal time . the trailing edge , however , is advanced by t j . this data - dependent jitter causes greater sensitivity to skew and jitter in the signal or sampling clock and may introduce noise into the timing loop . the waveform of fig4 c illustrates the situation when the signal is equalized by attenuating the dc and low frequency components so that all components are attenuated by a factor of a . here the height of the eye opening is a , considerably larger than 2a - 1 , especially for large attenuations . also , because the waveforms cross at the midpoint of their swing , the width of the eye is a full bit - cell giving better tolerance of timing skew and jitter . equalization eliminates the problem of frequency - dependent attenuation by filtering the transmitted or received waveform so the concatenation of the transmission line and the equalizing filter gives a flat frequency response . with equalization , an isolated 1 ( 0 ) in a field of 0s ( 1s ) crosses the receiver threshold at the midpoint of its swing , as shown in fig4 c , rather than being offset by an unattenuated dc component , as shown in fig4 a . narrow - band voice , video , and data modems have long used equalization to compensate for the linear portion of the line characteristics ( lee , edward a ., and messerschmitt , david g ., digital communication , second edition , kluwer , 1994 ). dally first used pre - emphasis equalization in broadband signaling with a wide bandwidth ( i . e ., greater than 100 mhz ) over short distances . in an embodiment of the present invention , the line is equalized using an analog fir filter built at the receiver . preferred implementations of the invention include analog finite input response ( fir ) filters , and fig5 and 6 illustrate one such implementation . the present invention comprises an equalizer that compensates for the low - pass characteristics of a communication channel , such as cable , board traces , etc ., in a multi - gbps link . a receiver uses this equalizer to cancel intersymbol interference ( isi ) caused by the channel . the equalizer uses an analog fir architecture that allows very fast processing speeds . filtering is performed in the analog domain directly on the analog sampled and held data values before they are digitized and used by other blocks . to implement an n - tap analog fir filter according to the present invention , n analog samples of a received data stream are sampled by the receiver at least once every symbol period . in other words , for each received data symbol , its value and the previous n − 1 symbol values are sampled and held . the n samples are then modulated by appropriate constant values , i . e ., filter tap weights , and added or subtracted in the analog domain . this capability allows the implementation of the fir function as follows : s eq ( n )= s n ∓ αs n − 1 ∓ βs n − 2 ( eq . 1 ) where s eq ( n ) is the filter output corresponding to sample n ; s n , s n − 1 , s n − 2 , etc . are the sampled analog data values at instances n , n − 1 , n − 2 , etc . ; and α , β , etc . are the associated filter tap weights . in the case of oversampling , for example for purposes such as clock recovery , subsymbol - spaced fir filtering can also be implemented using this technique , which allows frequency compensation for a larger frequency range . as an example , fig5 shows the half - circuit architecture of a differential demultiplexing receiver using a 2 - tap symbol - space fir equalizer . using multiple clock phases , the receiver switch samplers provide the required analog voltage samples from present and previous bit times , which are next converted into proportional currents based on the filter tap weight . by holding the present and former differential current samples , the equalizer can subtract the weighted value of the former sample s n − 1 from the present sample s n . specifically , in fig5 , the differential input 10 of the received data stream is sampled by samplers 12 , each clocked by a separate phase of a sampling clock ck 0 , ck 1 , etc . as an example , the determination of the value of data bit d 1 is discussed . s 1 is the corresponding sampled value , while s 0 corresponds to the previous sample . the output of the previous sample s 0 is multiplied by the tap weight α by multiplier 18 . this product αs 0 is then subtracted from sample s 1 by adder 20 . the difference s eq = s 01 =( s 1 − αs 0 ) is then held by sampler 22 and sensed by a sense amp 24 . fig6 illustrates a particular circuit which performs the operation of the multiplier 18 and adder 20 of fig5 , collectively shown in box 16 . the filtering function is performed by current summing two differential values with opposite polarity . these are s n0 and s p0 , the differential values corresponding to s 0 of fig5 , and s n1 and s p1 , the differential values corresponding to s 1 of fig5 . the currents shown in fig5 are due only to the signal inputs . dc bias currents are not shown , but are well - known within the art . transistors q 1 and q 2 form a differential amplifier , having as input the differential signal s 1 , comprising s p1 and s n1 . looking at just one side of the amplifier , the current resulting from input signal s p1 is i 1 . a voltage related to the tap weight α is applied to the previous sample s 0 by the dual input differential amplifier comprising transistors q 3 – q 6 . transistors q 5 and q 6 operate in the triode region , acting like resistors to scale the output current of the circuit . again , looking at just one half of this circuit , the previous sample differential signal s n0 is applied to transistor q 3 , while the weight a is applied to transistor q 5 , operating in its linear region as a resistor . the result is the weighted current − αi 0 , where − i 0 is the current that would result for a weight of α = 1 . by hard - wiring together the n - side of one differential pair with the p - side of the other differential pair , i . e ., by connecting the output of transistor q 3 with the output of transistor q 1 , the weighed current αi 0 is subtracted from the current i 1 , yielding the difference i 01 = i 1 − αi 0 . drawn through a passive or active resistor 36 , current i 01 is converted to voltage s 0n1 , through passive or active resistor r 1 to form one half of the differential output s 0 . the other half of the output , s 0p1 , is similarly formed by the other sides of each differential pair and resistor r 2 . this technique has been successfully reduced to practice in a transceiver chip using 0 . 3 um cmos technology . this equalizer shows successful results in the link response at 6 gb / s , by improving the eye diagram width by 20 %. see ramin farjad , et al ., “ a 0 . 3 um cmos 8 - gbps 4 - pam serial link transceiver ,” ieee jssc , march 2000 . the value of the weight a depends on the type of transmission medium . it should always be below unity , and typically a reasonable value is between 0 . 1 and 0 . 7 . while this invention has been particularly shown and described with references to preferred embodiments thereof , it will be understood by those skilled in the art that various changes in form and details may be made therein without departing from the scope of the invention encompassed by the appended claims . in particular , although a 2 - tap filter appears to be sufficient , the present invention extends to n - tap filters where n can be any number .