fig1 shows that a signal frame 1 has a downlink phase 2 in which downlink signals ds and dc are transmitted from the base station via the wireless transmission channel to the subscribers , and an uplink phase 3 in which uplink signals uc and us are transmitted from subscribers to the base station via the transmission channel for communication between a base station and a plurality of communication subscribers , e . g ., in a mobile wireless network . fig1 shows short intervals 4 without transmission power prior to the beginning of downlink phase 2 and uplink phase 3 , resulting from the switchover of the devices between transmission and reception operation ( transceiver turnaround interval ). these intervals without transmission power cause the zero signals used as frame synchronization signals to become ambiguous . the organization of the communication on the transmission channel takes place through the base station by transmitting downlink signal dc in downlink phase 2 through which the individual subscribers are assigned time slots for the transmission of data during uplink phase 3 . in uplink phase 3 the subscribers inform the base station of any intention to transmit , which is taken into account when assigning time slots during the next downlink phase 2 . fig2 shows details of the downlink signals in downlink phase 2 , which follows switchover interval 4 . the downlink signal in downlink period 2 begins with a frame synchronization signal 5 , which may be configured according to the present invention as explained in greater detail below with reference to the embodiments . a preamble 6 and two ofdm symbols 7 for fine synchronization follow . these are followed by the required number of data packets 8 , from which the variable length of downlink phase 2 results . one embodiment of a frame synchronization signal 5 is shown in fig3 . n frequencies for subcarriers of an ofdm signal in discrete equal intervals , of which n / 2 subcarriers are located on the positive side and n / 2 subcarriers are located on the negative side of a main carrier frequency , are plotted on the abscissa . in the embodiment illustrated , the n / 2 subcarriers on the negative ( n ) side are turned off , i . e ., have no signal power . however , the n / 2 subcarriers on the positive ( p ) side are transmitted with the same amplitude to form the frame synchronization signal . fig4 shows the basic structure of a receiver for an ofdm received signal . the received signal is amplified in an amplifier 10 and pre - filtered using a band - pass filter 11 . in a mixer 12 , which receives a first reference frequency f ref1 , the received signal is mixed down to an intermediary frequency , then filtered again using a band - pass filter 13 and split in a branch feeder unit 14 . an output branch of branch feeder unit 14 is connected to the input of another mixer 15 , which receives a second reference signal f ref2 . the signal thus mixed into the baseband goes , via an analog - digital converter 16 , to an ofdm demodulator 17 , as well as to a synchronization unit 18 . synchronization unit 18 is responsible for block , clock , and frequency synchronization . the functions of ofdm demodulator 17 and synchronization unit 18 are controlled by a controller unit 19 . the other output of branch feeder unit 14 goes to a detection device 20 to detect the frame synchronization signal . detection device 20 generates an output signal with which controller unit 19 is informed of the fact and the time of occurrence of a frame symbol . subsequently controller unit 19 issues a command to synchronization unit 18 to perform exact synchronization . when this has been completed , synchronization unit 18 transfers the data obtained to ofdm demodulator 17 via frequency offset and block begin ; ofdm demodulator 17 then demodulates the signal and provides a received data sequence at the output . the analog implementation of the frame synchronization signal detection illustrated here has the advantage that it can be performed completely independently of the remaining digital signal processing . the frame start is not detected by actively observing the channel ( as is the case with correlation ). instead , upon occurrence of the frame symbol , an event is triggered , i . e ., the receiver is passive and is informed by the frame synchronization signal . thus the method according to the present invention can also be used for waking up the receiver from a power - saving mode , whereby energy - efficient mobile subscriber terminals can be implemented . fig5 shows the design of a detection device 20 which receives the branched intermediary frequency signal at the output of branch feeder unit 14 . this input signal is split into two branches which have a band pass filter 21 , 22 , a downstream squaring element 23 , 24 , and a downstream low - pass filter 25 , 26 , respectively . band pass filter 21 filters out the upper frequency band ( p ) and band pass filter 22 filters out the lower frequency band ( n ). both filtered - out signal components are mixed non - linearly into the baseband by squaring element 23 , 24 and filtered by a low - pass filter 25 , 26 . the resulting signal in either branch is proportional to the received power within a time window to , which can be varied via the bandwidth of low - pass filter 25 , 26 . signals s p ( t ) and s n ( t ) thus formed are compared in a comparator 27 . a sufficiently large difference indicates the reception of the frame synchronization signal . detection device 20 ′ illustrated in fig6 has the identical components 21 through 26 for generating signals s p ( t ) and s n ( t ), as well as an identical comparator 27 for the frame synchronization signal according to fig3 . in addition , an adder 28 is also provided in which the two signals s p ( t ) and s n ( t ) are added and are detected in an analyzer 29 for detecting a zero signal . a sum at the output of adder 28 that is smaller than a threshold value slightly exceeding the noise intensity indicates a zero signal . when a frame synchronization signal is detected , comparator 27 outputs a positive output signal ; if a zero signal is detected , analyzer 29 outputs a positive output signal . on the basis of a suitable delay of one of the output signals , an and gate 30 may output a frame detection signal d ( t ) to controller 19 . by the use of detection criteria thus implemented for the beginning of a signal frame 1 , the likelihood of erroneous detection is considerably reduced . the necessary condition is , of course , that the respective transmitter should transmit a zero signal at the beginning frame 1 immediately before or after the frame synchronization signal according to fig3 . comparator 27 compares signals s p ( t ) and s n ( t ) and generates an appropriate output signal d ( t ). d ⁡ ( t ) = { ⁢ 1 if ⁢ ⁢ s p ⁡ ( t ) - s n ⁡ ( t ) & gt ; threshold 0 otherwise ⁢ for this purpose the inverted signal s n ( t ) is sent to an adder 30 and the difference thus formed is compared to a set threshold in a threshold value detector 31 . if the difference exceeds the set threshold of threshold value detector 31 , a potential characterizing the detection of the frame synchronization signal is generated at the output of threshold value detector 31 . this arrangement has the lowest degree of complexity , but has the disadvantage that the optimum threshold depends on the damping of the transmission signal . this disadvantage can be avoided using the embodiment according to fig8 , in which the two power signals s p ( t ) and s n ( t ) are divided by each other . therefore the condition for output signal d ( t ) is d ⁡ ( t ) = { ⁢ 1 if ⁢ ⁢ log ( s p ⁡ ( t ) s n ⁡ ( t ) ) & gt ; threshold 0 otherwise ⁢ hardware implementation includes sending input signals s p ( t ) and s n ( t ) to a log module 32 , 33 , respectively , and then forming the difference of the logged signals in adder 30 . mathematically this corresponds to forming the log of the quotients of signals s p ( t ) and s n ( t ). fig9 shows a comparator 27 working with the condition d ⁡ ( t ) = { ⁢ 1 if ⁢ ⁢ log ( s p ⁡ ( t ) - s n ⁡ ( t ) s p ⁡ ( t ) + s n ⁡ ( t ) ) & gt ; threshold 0 otherwise for this purpose log modules 32 , 33 have an upstream adder 34 , 35 , respectively , adder 34 receiving input signal s p ( t ) and inverted input signal s n ( t ), and adder 35 receiving input signals s p ( t ) and s n ( t ) without inversion . this design of comparator 27 allows the variance of the time of detection to be reduced compared to the method of fig8 . fig1 shows a variant of fig3 for the configuration of the frame synchronization signal according to the present invention . also in this case , the amplitudes of the subcarriers of the negative side ( n ) are zero . however , on the positive ( p ) side only every second subcarrier carries transmission power , while the subcarriers in between also have a zero amplitude . this results in a periodic frame symbol in the time range . this periodicity can be analyzed using a correlation and used for fine synchronization in synchronization unit 18 . the frame synchronization signal according to fig1 is also useful for performing digital frame detection . the received signal is sampled and processed using a fourier transform ( fft ), with an fft window of the one - half symbol length ( n / 2 ) being used . this guarantees that at least one symbol section is free of interblock interference for each transmitted ofdm symbol . by occupying only every other subcarrier within the positive ( p ) or negative ( n ) frequency range the spectrum of absolute values of the frame symbol becomes independent of the position of the time window . a frame synchronization signal is detected exactly at block k , at which the variable d k = ( ρ k - n k ) ⁢ ( ρ k + n k ) ρ k · n k ⁢ ⁢ where ρ k = ∑ i = 1 n / 2 ⁢ ⁢  r i . k  2 ⁢ ⁢ and ⁢ ⁢ n k = ∑ i = - 1 - n / 2 ⁢ ⁢  r i . k  2 exceeds a predefined threshold value . r i , k denotes the complex output signal of the n / 2 fft at the frequency iδf ( δf = subcarrier interval ) and the time kt / 2 ( t = useful duration of an ofdm symbol ). the frame synchronization method according to the present invention is particularly well - suited for ofdm transmission which is also presented in the example , because in ofdm it is much simpler to generate signals than in the single - carrier method . however the method according to the present invention can also be used in principle with single - carrier transmission methods . in this case sampling values of the time signal can be stored and read when needed . the intermediary frequency used in the receiver should be as low as possible . this ensures that the bandpass filters can be implemented in a simple manner . although the frame synchronization signals illustrated in fig3 and 10 have only one occupied subband ( p ) and one unoccupied subband ( n ), it is also possible in principle to divide the spectrum into four parts , for example , and have different occupancies in the subbands . further subdivision of the spectrum is possible ; however , in general , it is not considered practical . when the frame synchronization signal only contains power components in a subband , it can still be used as a reference signal for setting an amplitude amplification in the receiver . for this purpose , the frame synchronization signal should be selected so that the envelope is as constant as possible in the time range . this is important for using the frame synchronization signal for setting the amplification control and also to avoid overdriving the transmission amplifier . the occupancy of the individual subcarriers for the frame synchronization signal is independent of the type of modulation because the frame synchronization signal is not demodulated . therefore , any points in the signal space ( complex plane ) can be selected . in certain cases it has also been found useful to ensure the detection of uplink phase 3 . one possible method is switching around the positive and negative side band . for example , only the positive side band ( p ) may be occupied to detect the beginning of a downlink phase 2 , while only the negative side band ( n ) is occupied to mark the beginning of uplink phase 3 . for this purpose one of the comparisons performed by comparator 27 may take place using log modules . only the signal applied to threshold detector 31 changes . in order to reduce the likelihood of erroneous detection , as a variation to the arrangement of fig6 , instead of detecting a zero signal and a frame synchronization signal , a combination of frame synchronization signal 1 — frame synchronization signal 2 — can also be used to reduce the likelihood of erroneous detection . in this case , for example , all positive subcarrier frequencies may be occupied for the first frame detection signal and all negative subcarrier frequencies may be occupied for the second frame detection signal . the method according to the present invention has been described for use in time division duplex ( tdd ) systems . it is , however , also possible to use the method in frequency division duplex ( fdd ) systems with the appropriate modifications . this can be advantageous , in particular , for implementing power - saving subscriber terminals .